Product Sample & Technical Tools & Support & Folder Buy Documents Software Community TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 TPS54060A 0.5-A, 60-V Step-Down DC–DC Converter With Eco-Mode™ 1 Features 1 • 3.5Vto60VInputVoltageRange 3 Description • 200-mΩ High-SideMOSFET The TPS54060A device is a 60V, 0.5A, step down regulator with an integrated high side MOSFET. • HighEfficiencyatLightLoadswithaPulse Current mode control provides simple external SkippingEco-Mode™ compensation and flexible component selection. A • TighterEnableThresholdthanTPS54060 low ripple pulse skip mode reduces the no load, forMoreAccurateUVLOVoltage regulated output supply current to 116μA. Using the • AdjustableUVLOVoltageandHysteresis enable pin, shutdown supply current is reduced to 1.3μA,whentheenablepinislow. • 116μAOperatingQuiescentCurrent • 1.3μAShutdownCurrent Under voltage lockout is internally set at 2.5V, but can be increased using the enable pin. The output • 100kHzto2.5MHzSwitchingFrequency voltage startup ramp is controlled by the slow start • SynchronizestoExternalClock pin that can also be configured for • AdjustableSlowStart/Sequencing sequencing/tracking. An open drain power good signal indicates the output is within 94% to 107% of • UVandOVPowerGoodOutput itsnominalvoltage. • 0.8-VInternalVoltageReference A wide switching frequency range allows efficiency • MSOP-10and3mmx3mmVSON-10Package and external component size to be optimized. With PowerPAD™ Frequency fold back and thermal shutdown protects • Supportedby WEBENCH®andSwitcherPro™ thepartduringanoverloadcondition. SoftwareTool DeviceInformation(1) 2 Applications PARTNUMBER PACKAGE BODYSIZE(NOM) • 12-V,24-Vand48-VIndustrialandCommercial HVSSOP(10) 3.00mm×3.00mm TPS54060A LowPowerSystems VSON(10) 3.00mm×3.00mm • AftermarketAutoAccessories:Video,GPS, (1) For all available packages, see the orderable addendum at Entertainment theendofthedatasheet. SimplifiedSchematic EfficiencyvsLoadCurrent 100 V VVIINN PWRGD IN 90 TPS54060A 80 70 EN BOOT %) 60 SS/TR PH VOUT Efficiency ( 4500 RT/CLK 30 COMP 20 VI= 12 V VSENSE VO= 3.3 V 10 ƒsw= 500 kHz 0 GND 0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5 Load Current (A) C033 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectualpropertymattersandotherimportantdisclaimers.PRODUCTIONDATA.

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Table of Contents 1 Features.................................................................. 1 7.4 DeviceFunctionalModes........................................27 2 Applications........................................................... 1 8 ApplicationandImplementation........................ 29 3 Description............................................................. 1 8.1 ApplicationInformation............................................29 4 RevisionHistory..................................................... 2 8.2 TypicalApplication .................................................29 8.3 SystemExamples...................................................38 5 PinConfigurationandFunctions......................... 3 9 PowerSupplyRecommendations...................... 40 6 Specifications......................................................... 4 10 Layout................................................................... 40 6.1 AbsoluteMaximumRatings .....................................4 6.2 ESDRatings..............................................................4 10.1 LayoutGuidelines.................................................40 6.3 RecommendedOperatingConditions.......................4 10.2 LayoutExample....................................................41 6.4 ThermalInformation..................................................5 11 DeviceandDocumentationSupport................. 42 6.5 ElectricalCharacteristics...........................................5 11.1 DeviceSupport......................................................42 6.6 TypicalCharacteristics..............................................7 11.2 CommunityResources..........................................42 7 DetailedDescription............................................ 11 11.3 Trademarks...........................................................42 7.1 Overview.................................................................11 11.4 ElectrostaticDischargeCaution............................42 7.2 FunctionalBlockDiagram.......................................12 11.5 Glossary................................................................42 7.3 FeatureDescription.................................................12 12 Mechanical,Packaging,andOrderable Information........................................................... 42 4 Revision History NOTE:Pagenumbersforpreviousrevisionsmaydifferfrompagenumbersinthecurrentversion. ChangesfromRevisionB(September2013)toRevisionC Page • AddedPinConfigurationandFunctionssection,ESDRatingstable,FeatureDescriptionsection,DeviceFunctional Modes,ApplicationandImplementationsection,PowerSupplyRecommendationssection,Layoutsection,Device andDocumentationSupportsection,andMechanical,Packaging,andOrderableInformationsection .............................. 1 • RemovedOrderingInformationtable .................................................................................................................................... 1 ChangesfromRevisionA(September2012)toRevisionB Page • DeletedSWIFTfromthedatasheetTitleandFeatures......................................................................................................... 1 ChangesfromOriginal(March2012)toRevisionA Page • ChangedinDESCRIPTIONsecondparagraph93%to94%................................................................................................. 1 • Deletedtheword'output'inthePINFUNCTIONStable,BOOTrow,Descriptioncol........................................................... 3 • DeletedthesecondrowintheThermalInformationtable...................................................................................................... 4 • ChangedintheABSMAXTABLE,rowElecDischarge(HBM),maxvaluefrom1to2 ....................................................... 4 • ChangedthevaluesoftheHysteresiscurrentintheElectricalCharacteristicstable............................................................ 5 • ChangedintheELECCHARAtable,Timing....CLOCKsection,MINcol0.45to0.5............................................................ 6 • AddedparagraphandFigure29regardingnodevoltage.................................................................................................... 15 • Changedpartofsecondparagraphunderequation6fromwhentheVINUVLOisexceeded,TO:thevoltageatthe VINpinisbelowtheVINUVLO............................................................................................................................................ 16 • ChangedintheAPPLICATIONINFORMATIONsection,firsttable,1.5Ato0.5A............................................................... 29 • ChangedVOUTfrom5.5Vto3.3VinFigure51................................................................................................................... 30 • Changedinparagraphunderequation44:92uA/Vto97uA/V............................................................................................ 35 • Changedunderequation52Where:IOUTto:Io................................................................................................................. 35 2 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 5 Pin Configuration and Functions DGQandDRCPackage 10-PinVSONandMSOP TopView BOOT 1 10 PH BOOT 1 10 PH VIN 2 9 GND VIN 2 9 GND Thermal Thermal EN 3 8 COMP EN 3 Pad 8 COMP Pad (11) (11) SS/TR 4 7 VSENSE SS/TR 4 7 VSENSE RT/CLK 5 6 PWRGD RT/CLK 5 6 PWRGD PinFunctions PIN I/O DESCRIPTION NAME NO. AbootstrapcapacitorisrequiredbetweenBOOTandPH.Ifthevoltageonthiscapacitorisbelowthe BOOT 1 O minimumrequiredbythedevice,theoutputisforcedtoswitchoffuntilthecapacitorisrefreshed. Erroramplifieroutput,andinputtotheoutputswitchcurrentcomparator.Connectfrequencycompensation COMP 8 O componentstothispin. Enablepin,internalpull-upcurrentsource.Pullbelow1.2Vtodisable.Floattoenable.Adjusttheinput EN 3 I undervoltagelockoutwithtworesistors. GND 9 – Ground PH 10 I Thesourceoftheinternalhigh-sidepowerMOSFET. THERMAL GNDpinmustbeelectricallyconnectedtotheexposedpadontheprintedcircuitboardforproper 11 – PAD operation. Anopendrainoutput,assertslowifoutputvoltageislowduetothermalshutdown,dropout,over-voltageor PWRGD 6 O ENshutdown. ResistorTimingandExternalClock.Aninternalamplifierholdsthispinatafixedvoltagewhenusingan externalresistortogroundtosettheswitchingfrequency.IfthepinispulledabovethePLLupper RT/CLK 5 I threshold,amodechangeoccursandthepinbecomesasynchronizationinput.Theinternalamplifieris disabledandthepinisahighimpedanceclockinputtotheinternalPLL.Ifclockingedgesstop,theinternal amplifierisre-enabledandthemodereturnstoaresistorsetfunction. Slow-startandTracking.Anexternalcapacitorconnectedtothispinsetstheoutputrisetime.Sincethe SS/TR 4 I voltageonthispinoverridestheinternalreference,itcanbeusedfortrackingandsequencing. VIN 2 I Inputsupplyvoltage,3.5Vto60V. VSENSE 7 I Invertingnodeofthetransconductance(gm)erroramplifier. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 3 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings(1) overoperatingtemperaturerange(unlessotherwisenoted). MIN MAX UNIT VIN –0.3 65 EN –0.3 5 BOOT 73 VSENSE –0.3 3 Inputvoltage V COMP –0.3 3 PWRGD –0.3 6 SS/TR –0.3 3 RT/CLK –0.3 3.6 BOOT-PH 8 Outputvoltage PH –0.6 65 V PH,10-nsTransient –2 65 VoltageDifference PADtoGND ±200 mV EN 100 μA BOOT 100 mA Sourcecurrent VSENSE 10 μA PH CurrentLimit A RT/CLK 100 μA VIN CurrentLimit A COMP 100 μA Sinkcurrent PWRGD 10 mA SS/TR 200 μA Operatingjunctiontemperature –40 150 °C Storagetemperature –65 150 °C (1) Stressesbeyondthoselistedunderabsolutemaximumratingsmaycausepermanentdamagetothedevice.Thesearestressratings onlyandfunctionaloperationofthedeviceattheseoranyotherconditionsbeyondthoseindicatedunderrecommendedoperating conditionsRecommendedOperatingConditionsisnotimplied.Exposuretoabsolute-maximum-ratedconditionsforextendedperiods mayaffectdevicereliability. 6.2 ESD Ratings VALUE UNIT Humanbodymodel(HBM),perANSI/ESDA/JEDECJS-001(1)QSS009- ±2000 105(JESD22-A114A) V Electrostaticdischarge V (ESD) Charged-devicemodel(CDM),perJEDECspecificationJESD22- C101(2)QSS009-147(JESD22-C101B.01) ±500 (1) JEDECdocumentJEP155statesthat500-VHBMallowssafemanufacturingwithastandardESDcontrolprocess.. (2) JEDECdocumentJEP157statesthat250-VCDMallowssafemanufacturingwithastandardESDcontrolprocess.. 6.3 Recommended Operating Conditions overoperatingfree-airtemperaturerange(unlessotherwisenoted) MIN MAX UNIT V Supplyinputvoltagerange 3.5 60 V IN V Outputvoltagerange 0.8 58 V O I Outputcurrentrange 0 0.5 A O T JunctionTemperature –40 150 °C J 4 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 6.4 Thermal Information TPS54060A THERMALMETRIC(1) DGQ(HVSSOP) DRC(VSON) UNIT 10PINS 10PINS R Junction-to-ambientthermalresistance 62.5 40 °C/W θJA R Junction-to-case(top)thermalresistance 1.7 0.6 °C/W θJC(top) R Junction-to-boardthermalresistance 20.1 7.5 °C/W θJB ψ Junction-to-topcharacterizationparameter 83 65 °C/W JT ψ Junction-to-boardcharacterizationparameter 21 7.8 °C/W JB R Junction-to-case(bottom)thermalresistance 28 8 °C/W θJC(bot) (1) Formoreinformationabouttraditionalandnewthermalmetrics,seetheSemiconductorandICPackageThermalMetricsapplication report,SPRA953. 6.5 Electrical Characteristics T =–40°Cto150°C,VIN=3.5to60V(unlessotherwisenoted) J PARAMETER TESTCONDITIONS MIN TYP MAX UNIT SUPPLYVOLTAGE(VINPIN) Operatinginputvoltage 3.5 60 V Internalundervoltagelockout Novoltagehysteresis,risingandfalling 2.5 V threshold Shutdownsupplycurrent EN=0V,25°C,3.5V≤VIN≤60V 1.3 4 Operating:nonswitchingsupply μA VSENSE=0.83V,VIN=12V,25°C 116 136 current ENABLEANDUVLO(ENPIN) Enablethresholdvoltage Novoltagehysteresis,risingandfalling,25°C 1.11 1.25 1.36 V Enablethreshold+50mV –3.8 Inputcurrent μA Enablethreshold–50mV –0.9 Hysteresiscurrent 1.91 2.95 3.99 μA VOLTAGEREFERENCE T =25°C 0.792 0.8 0.808 J Voltagereference V 0.784 0.8 0.816 HIGH-SIDEMOSFET VIN=3.5V,BOOT-PH=3V 300 On-resistance mΩ VIN=12V,BOOT-PH=6V 200 410 ERRORAMPLIFIER Inputcurrent 50 nA Erroramplifiertransconductance(g ) –2μA<I <2μA,V =1V 97 μMhos M COMP COMP Erroramplifiertransconductance(gM) –2μA<ICOMP<2μA,VCOMP=1V, 26 μMhos duringslowstart V =0.4V VSENSE Erroramplifierdcgain V =0.8V 10,000 V/V VSENSE Erroramplifierbandwidth 2700 kHz Erroramplifiersource/sink V =1V,100mVoverdrive ±7 μA (COMP) COMPtoswitchcurrent 1.9 A/V transconductance CURRENTLIMIT Currentlimitthreshold VIN=12V,T =25°C 0.6 0.94 A J THERMALSHUTDOWN Thermalshutdown 182 °C TIMINGRESISTORANDEXTERNALCLOCK(RT/CLKPIN) SwitchingFrequencyRangeusing 100 2500 kHz RTmode Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 5 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Electrical Characteristics (continued) T =–40°Cto150°C,VIN=3.5to60V(unlessotherwisenoted) J PARAMETER TESTCONDITIONS MIN TYP MAX UNIT f Switchingfrequency R =200kΩ 450 581 720 kHz SW T SwitchingFrequencyRangeusing 300 2200 kHz CLKmode MinimumCLKinputpulsewidth 40 ns RT/CLKhighthreshold 1.9 2.2 V RT/CLKlowthreshold 0.5 0.7 V RT/CLKfallingedgetoPHrising Measuredat500kHzwithRTresistorinseries 60 ns edgedelay PLLlockintime Measuredat500kHz 100 μs SLOWSTARTANDTRACKING(SS/TR) Chargecurrent V =0.4V 2 μA SS/TR SS/TR-to-VSENSEmatching V =0.4V 45 mV SS/TR SS/TR-to-referencecrossover 98%nominal 1.0 V SS/TRdischargecurrent(overload) VSENSE=0V,V(SS/TR)=0.4V 112 μA SS/TRdischargevoltage VSENSE=0V 54 mV POWERGOOD(PWRGDPIN) VSENSEfalling 92% VSENSErising 94% V VSENSEthreshold VSENSE VSENSErising 109% VSENSEfalling 107% Hysteresis VSENSEfalling 2% Outputhighleakage VSENSE=VREF,V(PWRGD)=5.5V,25°C 10 nA Onresistance I(PWRGD)=3mA,VSENSE<0.79V 50 Ω MinimumVINfordefinedoutput V(PWRGD)<0.5V,II(PWRGD)=100μA 0.95 1.5 V 6 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 6.6 Typical Characteristics W)500 0.816 m e ( nc On-State Resista327550 BOOT-PH = 3 VBOOT-PH = 6 V Reference (V) 00..880080 n-Source 125 Voltage 0.792 ai Dr atic St 0 0.784 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C001 C002 Figure1.OnResistancevsJunctionTemperature Figure2.VoltageReferencevsJunctionTemperature 1.1 610 600 1 Hz) Switch Current (A) 00..89 witching Frequency (k555789000 S 560 0.7 550 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C003 C004 Figure3.SwitchCurrentLimitvsJunctionTemperature Figure4.SwitchingFrequencyvsJunctionTemperature 2500 500 2000 400 Hz) Hz) witching Frequency (k11050000 witching Frequency (k 230000 S 500 S 100 0 0 0 25 50 75 100 125 150 175 200 200 300 400 500 600 700 800 900 1000 1100 1200 RT/CLK Resistance (kW) RT/CLK Resistance (kW) C005 C006 Figure5.SwitchingFrequencyvsRT/CLKResistanceHigh Figure6.SwitchingFrequencyvsRT/CLKResistanceLow FrequencyRange FrequencyRange Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 7 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Typical Characteristics (continued) 40 150 130 30 V) V)110 A/ A/ μ μ m ( m ( g g 90 20 70 10 50 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C007 C008 Figure7.EATransconductanceDuringSlowStartvs Figure8.EATransconductancevsJunctionTemperature JunctionTemperature 1.40 –3.25 –3.5 V) 1.30 d ( ol A) Thresh μI((EN)–3.75 N E 1.20 –4 1.10 –4.25 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C009 C010 Figure9.ENPinVoltagevsJunctionTemperature Figure10.ENPinCurrentvsJunctionTemperature –0.8 –1 –0.85 –1.5 A) A) μI((EN) –0.9 μI((SS/TR) –2 –0.95 –2.5 –1 –3 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C011 C012 Figure11.ENPinCurrentvsJunctionTemperature Figure12.SS/TRChargeCurrentvsJunctionTemperature 8 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Typical Characteristics (continued) 120 100 80 115 A) nal fsw 60 μI(I(SS/TR)110 of Nomi 40 % 105 20 100 0 –50 –25 0 25 50 75 100 125 150 0 0.2 0.4 0.6 0.8 JunctionTemperature (°C) V (V) C013 SENSE C014 Figure13.SS/TRDischargeCurrentvsJunction Figure14.SwitchingFrequencyvsVSENSE Temperature 2 2 1.5 1.5 A) A) μ μ ( 1 ( 1 I(VIN) I(VIN) 0.5 0.5 0 0 –50 –25 0 25 50 75 100 125 150 0 10 20 30 40 50 60 JunctionTemperature (°C) Input Voltage (V) C015 C016 Figure15.ShutdownSupplyCurrentvsJunction Figure16.ShutdownSupplyCurrentvsInputVoltage(V ) in Temperature 140 140 130 130 120 120 A) A) μ μ ( ( I(VIN)110 I(VIN)110 100 100 90 90 –50 –25 0 25 50 75 100 125 150 0 20 40 60 JunctionTemperature (°C) Input Voltage (V) C017 C018 Figure17.VINSupplyCurrentvsJunctionTemperature Figure18.VINSupplyCurrentvsInputVoltage Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 9 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Typical Characteristics (continued) 100 115 VSENSE Rising 80 V)ref110 % of 105 VSENSE Falling W) 60 d ( ON ( shol100 RDS 40 DThre 95 VSENSE Rising G R W 20 P 90 VSENSE Falling 0 85 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C019 C020 Figure19.PWRGDonResistancevsJunctionTemperature Figure20.PWRGDThresholdvsJunctionTemperature 2.5 3 2.25 2.75 V) ( V) VI(BOOT-PH) 2 V(I(VIN) 2.50 1.75 2.25 1.5 2 –50 –25 0 25 50 75 100 125 150 -50 -25 0 25 50 75 100 125 150 JunctionTemperature (°C) JunctionTemperature (°C) C021 C022 Figure21.BOOT-PHUVLOvsJunctionTemperature Figure22.InputVoltage(UVLO)vsJunctionTemperature 500 60 55 400 50 V)300 V) m m et ( et (45 s s Off200 Off 40 100 35 0 30 0 100 200 300 400 500 600 700 800 –50 –25 0 25 50 75 100 125 150 VSENSE (mV) JunctionTemperature (°C) C023 C024 Figure23.SS/TRToVSENSEOffsetvsVSENSE Figure24.SS/TRToVSENSEOffsetvsTemperature 10 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 7 Detailed Description 7.1 Overview The TPS54060A device is a 60-V, 0.5-A, step-down (buck) regulator with an integrated high side n-channel MOSFET. To improve performance during line and load transients the device implements a constant frequency, current mode control which reduces output capacitance and simplifies external frequency compensation design. The wide switching frequency of 100kHz to 2500kHz allows for efficiency and size optimization when selecting the output filter components. The switching frequency is adjusted using a resistor to ground on the RT/CLK pin. The device has an internal phase lock loop (PLL) on the RT/CLK pin that is used to synchronize the power switchturnontoafallingedgeofanexternalsystemclock. The TPS54060A has a default start up voltage of approximately 2.5V. The EN pin has an internal pull-up current source that can be used to adjust the input voltage under voltage lockout (UVLO) threshold with two external resistors. In addition, the pull up current provides a default condition. When the EN pin is floating the device will operate. The operating current is 116μA when not switching and under no load. When the device is disabled, the supplycurrentis1.3μA. The integrated 200-mΩ high side MOSFET allows for high efficiency power supply designs capable of delivering 0.5 amperes of continuous current to a load. The TPS54060A reduces the external component count by integrating the boot recharge diode. The bias voltage for the integrated high side MOSFET is supplied by a capacitor on the BOOT to PH pin. The boot capacitor voltage is monitored by an UVLO circuit and will turn the highsideMOSFEToffwhenthebootvoltagefallsbelowapresetthreshold.TheTPS54060Acanoperateathigh duty cycles because of the boot UVLO. The output voltage can be stepped down to as low as the 0.8V reference. The TPS54060A has a power good comparator (PWRGD) which asserts when the regulated output voltage is less than 92% or greater than 109% of the nominal output voltage. The PWRGD pin is an open drain output which deasserts when the VSENSE pin voltage is between 94% and 107% of the nominal output voltage allowingthepintotransitionhighwhenapull-upresistorisused. The TPS54060A minimizes excessive output overvoltage (OV) transients by taking advantage of the OV power good comparator. When the OV comparator is activated, the high side MOSFET is turned off and masked from turningonuntiltheoutputvoltageislowerthan107%. The SS/TR (slow start/tracking) pin is used to minimize inrush currents or provide power supply sequencing during power up. A small value capacitor should be coupled to the pin to adjust the slow start time. A resistor dividercanbecoupledtothepinforcriticalpowersupplysequencingrequirements.TheSS/TRpinisdischarged before the output powers up. This discharging ensures a repeatable restart after an over-temperature fault, UVLOfaultoradisabledcondition. The TPS54060A, also, discharges the slow start capacitor during overload conditions with an overload recovery circuit. The overload recovery circuit will slow start the output from the fault voltage to the nominal regulation voltage once a fault condition is removed. A frequency foldback circuit reduces the switching frequency during startupandovercurrentfaultconditionstohelpcontroltheinductorcurrent. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 11 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 7.2 Functional Block Diagram PWRGD EN VIN 6 3 2 Shutdown Thermal Shutdown UVLO Enable UV Logic Comparator Shutdown Shutdown Logic OV Enable Threshold Boot Charge Voltage Minimum Boot Reference Clamp UVLO Current Pulse Sense ERROR Skip AMPLIFIER PWM VSENSE 7 Comparator 1 BOOT SS/TR 4 Logic And PWM Latch Shutdown Slope Compensation COMP 8 10PH 11 POWERPAD Frequency Shift Overload Maximum Recovery Clamp Oscillator 9 GND with PLL TPS54060ABlock Diagram 5 RT/CLK 7.3 Feature Description 7.3.1 FixedFrequencyPWMControl The TPS54060A uses an adjustable fixed frequency, peak current mode control. The output voltage is compared through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives the COMP pin. An internal oscillator initiates the turn on of the high side power switch. The error amplifier output is compared to the high side power switch current. When the power switch current reaches the level set by the COMP voltage, the power switch is turned off. The COMP pin voltage will increase and decrease as the output current increases and decreases. The device implements a current limit by clamping the COMP pin voltage to a maximumlevel.TheEco-Mode™isimplementedwithaminimumclampontheCOMPpin. 7.3.2 SlopeCompensationOutputCurrent The TPS54060A adds a compensating ramp to the switch current signal. This slope compensation prevents sub- harmonicoscillations.Theavailablepeakinductorcurrentremainsconstantoverthefulldutycyclerange. 12 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) 7.3.3 BootstrapVoltage(BOOT) The TPS54060A has an integrated boot regulator, and requires a small ceramic capacitor between the BOOT and PH pins to provide the gate drive voltage for the high side MOSFET. The BOOT capacitor is refreshed when the high side MOSFET is off and the low side diode conducts. The value of this ceramic capacitor should be 0.1μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10V or higher is recommendedbecauseofthestablecharacteristicsovertemperatureandvoltage. To improve drop out, the TPS54060A is designed to operate at 100% duty cycle as long as the BOOT to PH pin voltage is greater than 2.1V. When the voltage from BOOT to PH drops below 2.1V, the high side MOSFET is turned off using an UVLO circuit which allows the low side diode to conduct and refresh the charge on the BOOT capacitor. Since the supply current sourced from the BOOT capacitor is low, the high side MOSFET can remain on for more switching cycles than are required to refresh the capacitor, thus the effective duty cycle of the switchingregulatorishigh. 7.3.4 LowDropoutOperation The effective duty cycle during dropout of the regulator is mainly influenced by the voltage drops across the power MOSFET, inductor resistance, low side diode and printed circuit board resistance. During operating conditions in which the input voltage drops and the regulator is operating in continuous conduction mode, the high side MOSFET can remain on for 100% of the duty cycle to maintain output regulation, until the BOOT to PH voltagefallsbelow2.1V. Attention must be taken in maximum duty cycle applications which experience extended time periods with light loads or no load. When the voltage across the BOOT capacitor falls below the 2.1V UVLO threshold, the high sideMOSFETisturnedoff,buttheremaynotbeenoughinductorcurrenttopullthePHpindowntorechargethe BOOT capacitor. The high side MOSFET of the regulator stops switching because the voltage across the BOOT capacitor is less than 2.1V. The output capacitor then decays until the difference in the input voltage and output voltage is greater than 2.1V, at which point the BOOT UVLO threshold is exceeded, and the device starts switching again until the desired output voltage is reached. This operating condition persists until the input voltage and/or the load current increases. It is recommended to adjust the VIN stop voltage greater than the BOOTUVLOtriggerconditionattheminimumloadoftheapplicationusingtheadjustableVINUVLOfeaturewith resistorsontheENpin. The start and stop voltages for typical 3.3V and 5V output applications are shown in Figure 25 and Figure 26. The voltages are plotted versus load current. The start voltage is defined as the input voltage needed to regulate the output within 1%. The stop voltage is defined as the input voltage at which the output drops by 5% or stops switching. During high duty cycle conditions, the inductor current ripple increases while the BOOT capacitor is being recharged resulting in an increase in ripple voltage on the output. This is due to the recharge time of the boot capacitorbeinglongerthanthetypicalhighsideofftimewhenswitchingoccurseverycycle. 4 5.6 3.8 5.4 V) V) ge ( 3.6 Start ge ( 5.2 Start a a olt olt V V ut 3.4 Stop ut 5 p p n n I I Stop 3.2 4.8 3 4.6 0 0.05 0.10 0.15 0.20 0 0.05 0.10 0.15 0.20 Output Current (A) Output Current (A) C025 C026 Figure25.3.3VStart/StopVoltage Figure26.5.0VStart/StopVoltage Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 13 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) 7.3.5 ErrorAmplifier The TPS54060A has a transconductance amplifier for the error amplifier. The error amplifier compares the VSENSE voltage to the lower of the SS/TR pin voltage or the internal 0.8V voltage reference. The transconductance (gm) of the error amplifier is 97μA/V during normal operation. During the slow start operation, the transconductance is a fraction of the normal operating gm. When the voltage of the VSENSE pin is below 0.8VandthedeviceisregulatingusingtheSS/TRvoltage,thegmis25μA/V. The frequency compensation components (capacitor, series resistor and capacitor) are added to the COMP pin toground. 7.3.6 VoltageReference The voltage reference system produces a precise ±2% voltage reference over temperature by scaling the output ofatemperaturestablebandgapcircuit. 7.3.7 AdjustingtheOutputVoltage The output voltage is set with a resistor divider from the output node to the VSENSE pin. It is recommended to use 1% tolerance or better divider resistors. Start with a 10 kΩ for the R2 resistor and use the Equation 1 to calculate R1. To improve efficiency at light loads consider using larger value resistors. If the values are too high the regulator will be more susceptible to noise and voltage errors from the VSENSE input current will be noticeable æVout - 0.8Vö R1=R2 ´ ç ÷ è 0.8V ø (1) 7.3.8 EnableandAdjustingUndervoltageLockout The TPS54060A is disabled when the VIN pin voltage falls below 2.5 V. If an application requires a higher undervoltage lockout (UVLO), use the EN pin as shown in Figure 27 to adjust the input voltage UVLO by using the two external resistors. Though it is not necessary to use the UVLO adjust registers, for operation it is highly recommended to provide consistent power up behavior. The EN pin has an internal pull-up current source, I1, of 0.9μA that provides the default condition of the TPS54060A operating when the EN pin floats. Once the EN pin voltage exceeds 1.25V, an additional 2.9μA of hysteresis, Ihys, is added. This additional current facilitates input voltage hysteresis. Use Equation 2 to set the external hysteresis for the input voltage. Use Equation 3 to set the inputstartvoltage. TPS54060A VIN Ihys I1 R1 0.9mA 2.9mA + R2 EN 1.25 V - Figure27. AdjustableUndervoltageLockout(UVLO) V -V R1= START STOP I HYS (2) V R2= ENA V -V START ENA +I R1 1 (3) 14 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) Another technique to add input voltage hysteresis is shown in Figure 28. This method may be used, if the resistance values are high from the previous method and a wider voltage hysteresis is needed. The resistor R3 sourcesadditionalhysteresiscurrentintotheENpin. TPS54060A VIN Ihys R1 I1 2.9mA 0.9mA + R2 EN 1.25 V - VOUT R3 Figure28. AddingAdditionalHysteresis V -V R1= START STOP V I + OUT HYS R3 (4) V R2= ENA V -V V START ENA +I - ENA R1 1 R3 (5) Do not place a low-impedance voltage source with greater than 5 V directly on the EN pin. Do not place a capacitor directly on the EN pin if V > 5 V when using a voltage divider to adjust the start and stop voltage. EN The node voltage, (see Figure 29) must remain equal to or less than 5.8 V. The zener diode can sink up to 100 µA. The EN pin voltage can be greater than 5 V if the V voltage source has a high impedance and does not IN sourcemorethan100µAintotheENpin. VIN R1 ENA Node 10kohm R2 5.8V Figure29. NodeVoltage Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 15 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) 7.3.9 SlowStart/TrackingPin(SS/TR) The TPS54060A effectively uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as the power-supply's reference voltage and regulates the output accordingly. A capacitor on the SS/TR pin to ground implements a slow start time. The TPS54060A has an internal pull-up current source of 2μA that charges the external slow start capacitor. The calculations for the slow start time (10% to 90%) are shown in Equation 6. The voltage reference (V ) is 0.8 V and the slow start current (I ) is 2μA. The slow start capacitor should REF SS remainlowerthan0.47μFandgreaterthan0.47nF. Tss(ms) ´ Iss(mA) Css(nF)= Vref(V) ´ 0.8 (6) At power up, the TPS54060A will not start switching until the slow start pin is discharged to less than 40 mV to ensureaproperpowerup,seeFigure30. Also, during normal operation, the TPS54060A will stop switching and the SS/TR must be discharged to 40 mV, the voltage at the VIN pin is below the VIN UVLO, EN pin pulled below 1.25V, or a thermal shutdown event occurs. The VSENSE voltage will follow the SS/TR pin voltage with a 45mV offset up to 85% of the internal voltage reference. When the SS/TR voltage is greater than 85% on the internal reference voltage the offset increases as the effective system reference transitions from the SS/TR voltage to the internal voltage reference (see Figure23).TheSS/TRvoltagewillramplinearlyuntilclampedat1.7V. EN SS/TR V SENSE VOUT Figure30. OperationofSS/TRPinwhenStarting 7.3.10 OverloadRecoveryCircuit The TPS54060A has an overload recovery (OLR) circuit. The OLR circuit will slow start the output from the overload voltage to the nominal regulation voltage once the fault condition is removed. The OLR circuit will discharge the SS/TR pin to a voltage slightly greater than the VSENSE pin voltage using an internal pull down of 100μA when the error amplifier is changed to a high voltage from a fault condition. When the fault condition is removed,theoutputwillslowstartfromthefaultvoltagetonominaloutputvoltage. 16 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) 7.3.11 Sequencing Many of the common power supply sequencing methods can be implemented using the SS/TR, EN and PWRGD pins. The sequential method can be implemented using an open drain output of a power on reset pin of another device. The sequential method is illustrated in Figure 31 using two TPS54060A devices. The power good is coupled to the EN pin on the TPS54060A which will enable the second power supply once the primary supply reaches regulation. If needed, a 1nF ceramic capacitor on the EN pin of the second power supply will provide a 1msstartupdelay.Figure32showstheresultsofFigure31. TPS54060A PWRGD EN EN EN1 SS/TR SS/TR PWRGD1 PWRGD VOUT1 VOUT2 Figure31.SchematicforSequentialStart-Up Figure32.SequentialStartupusingENand Sequence PWRGD TPS54060A 3 EN EN1, EN2 4 SS/TR 6 PWRGD VOUT1 TPS54060A VOUT2 3 EN 4 SS/TR 6 PWRGD Figure33.SchematicforRatiometricStart-Up Figure34.Ratio-MetricStartupusingCoupled Sequence SS/TRpins Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 17 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) Figure 33 shows a method for ratio-metric start up sequence by connecting the SS/TR pins together. The regulator outputs will ramp up and reach regulation at the same time. When calculating the slow start time the pullupcurrentsourcemustbedoubledinEquation6.Figure34showstheresultsofFigure33. TPS54060A EN VOUT1 SS/TR PWRGD TPS54060A EN VOUT2 R1 SS/TR R2 PWRGD R3 R4 Figure35. SchematicforRatiometricandSimultaneousStart-UpSequence Ratio-metric and simultaneous power supply sequencing can be implemented by connecting the resistor network of R1 and R2 shown in Figure 35 to the output of the power supply that needs to be tracked or another voltage reference source. Using Equation 7 and Equation 8, the tracking resistors can be calculated to initiate the Vout2 slightly before, after or at the same time as Vout1. Equation 9 is the voltage difference between Vout1 and Vout2 atthe95%ofnominaloutputregulation. The deltaV variable is zero volts for simultaneous sequencing. To minimize the effect of the inherent SS/TR to VSENSE offset (Vssoffset) in the slow start circuit and the offset created by the pullup current source (Iss) and trackingresistors,theVssoffsetandIssareincludedasvariablesintheequations. To design a ratio-metric start up in which the Vout2 voltage is slightly greater than the Vout1 voltage when Vout2 reachesregulation,useanegativenumberinEquation7throughEquation9fordeltaV.Equation9 willresultina positivenumberforapplicationswhichtheVout2isslightlylowerthanVout1whenVout2regulationisachieved. Since the SS/TR pin must be pulled below 40mV before starting after an EN, UVLO or thermal shutdown fault, careful selection of the tracking resistors is needed to ensure the device will restart after a fault. Make sure the calculated R1 value from Equation 7 is greater than the value calculated in Equation 10 to ensure the device can recoverfromafault. As the SS/TR voltage becomes more than 85% of the nominal reference voltage the Vssoffset becomes larger as the slow start circuits gradually handoff the regulation reference to the internal voltage reference. The SS/TR pin voltage needs to be greater than 1.3V for a complete handoff to the internal voltage reference as shown in Figure23. 18 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) Vout2+deltaV Vssoffset R1= ´ VREF Iss (7) VREF ´ R1 R2= Vout2+deltaV - VREF (8) deltaV=Vout1 - Vout2 (9) R1>2800 ´ Vout1 - 180 ´ deltaV (10) EN EN VOUT1 VOUT1 VOUT2 VOUT2 Figure36.Ratio-metricStartupwithTrackingResistors Figure37.RatiometricStartupwithTrackingResistors EN VOUT1 VOUT2 Figure38.SimultaneousStartupWithTrackingResistor Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 19 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) 7.3.12 ConstantSwitchingFrequencyandTimingResistor(RT/CLKPin) The switching frequency of the TPS54060A is adjustable over a wide range from approximately 100kHz to 2500kHz by placing a resistor on the RT/CLK pin. The RT/CLK pin voltage is typically 0.5V and must have a resistor to ground to set the switching frequency. To determine the timing resistance for a given switching frequency, use Equation 11 or the curves in Figure 39 or Figure 40. To reduce the solution size one would typically set the switching frequency as high as possible, but tradeoffs of the supply efficiency, maximum input voltageandminimumcontrollableontimeshouldbeconsidered. Theminimumcontrollableontimeistypically130nsandlimitsthemaximumoperatinginputvoltage. The maximum switching frequency is also limited by the frequency shift circuit. More discussion on the details of themaximumswitchingfrequencyislocatedbelow. 206033 RT (kW) = ¦sw (kHz)1.0888 (11) 2500 500 V= 12 V, I T = 25°C J Hz 2000 z) 400 k H Frequency - 1500 equency (k 300 witching 1000 ching Fr 200 f- Ss 500 Swit 100 0 0 0 25 50 75 100 125 150 175 200 200 300 400 500 600 700 800 900 1000 1100 1200 RT/CLK - Clock Resistance - kW RT/CLK Resistance (kW) Figure39.SwitchingFrequencyvsRT/CLKResistance C006 Figure40.SwitchingFrequencyvsRT/CLKResistance HighFrequencyRange LowFrequencyRange 7.3.13 OvercurrentProtectionandFrequencyShift The TPS54060A implements current mode control which uses the COMP pin voltage to turn off the high side MOSFET on a cycle by cycle basis. Each cycle the switch current and COMP pin voltage are compared, when the peak switch current intersects the COMP voltage, the high side switch is turned off. During overcurrent conditions that pull the output voltage low, the error amplifier will respond by driving the COMP pin high, increasing the switch current. The error amplifier output is clamped internally, which functions as a switch current limit. To increase the maximum operating switching frequency at high input voltages the TPS54060A implements a frequency shift. The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on VSENSEpin. The device implements a digital frequency shift to enable synchronizing to an external clock during normal startup and fault conditions. Since the device can only divide the switching frequency by 8, there is a maximum inputvoltagelimitinwhichthedeviceoperatesandstillhavefrequencyshiftprotection. Duringshort-circuitevents(particularlywithhighinputvoltageapplications),thecontrolloophasafiniteminimum controllable on time and the output has a low voltage. During the switch on time, the inductor current ramps to the peak current limit because of the high input voltage and minimum on time. During the switch off time, the inductor would normally not have enough off time and output voltage for the inductor to ramp down by the ramp upamount.Thefrequencyshifteffectivelyincreasestheofftimeallowingthecurrenttorampdown. 20 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) 7.3.14 SelectingtheSwitchingFrequency The switching frequency that is selected should be the lower value of the two equations, Equation 12 and Equation13.Equation12isthemaximumswitchingfrequencylimitationsetbytheminimumcontrollableontime. Settingtheswitchingfrequencyabovethisvaluewillcausetheregulatortoskipswitchingpulses. Equation 13 is the maximum switching frequency limit set by the frequency shift protection. To have adequate output short circuit protection at high input voltages, the switching frequency should be set to be less than the fsw(maxshift) frequency. In Equation 13, to calculate the maximum switching frequency one must take into accountthattheoutputvoltagedecreasesfromthenominalvoltageto0volts,thefdivintegerincreasesfrom1to 8correspondingtothefrequencyshift. In Figure 41, the solid line illustrates a typical safe operating area regarding frequency shift and assumes the output voltage is zero volts, and the resistance of the inductor is 0.130Ω, FET on resistance of 0.2Ω and the diode voltage drop is 0.5V. The dashed line is the maximum switching frequency to avoid pulse skipping. Enter these equations in a spreadsheet or other software or use the SwitcherPro design software to determine the switchingfrequency. fSW(maxskip)=æçt1 ö÷´æçç(I(LV´ -RIdc´+VROhUsT++VVdd))ö÷÷ è ONø è IN L ø (12) ¦div æ(I ´Rdc+V +Vd)ö L OUTSC ¦ = ´ç ÷ SW(shift) t ç (V -I ´Rhs+Vd) ÷ ON è IN L ø (13) I inductorcurrent L Rdc inductorresistance V maximuminputvoltage IN V outputvoltage OUT V outputvoltageduringshort OUTSC Vd diodevoltagedrop R switchonresistance DS(on) t controllableontime ON ƒ frequencydivideequals(1,2,4,or8) DIV 2500 2000 z) Shift H k y ( nc1500 e u q Fre Skip g 1000 n hi c wit S 500 0 10 20 30 40 50 60 Input Voltage (V) C027 Figure41. MaximumSwitchingFrequencyvs.InputVoltage Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 21 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) 7.3.15 HowtoInterfacetoRT/CLKPin The RT/CLK pin can be used to synchronize the regulator to an external system clock. To implement the synchronization feature connect a square wave to the RT/CLK pin through the circuit network shown in Figure 42. The square wave amplitude must transition lower than 0.5V and higher than 2.2V on the RT/CLK pin and have an on time greater than 40 ns and an off time greater than 40 ns. The synchronization frequency range is 300 kHz to 2200 kHz. The rising edge of the PH will be synchronized to the falling edge of RT/CLK pin signal. The external synchronization circuit should be designed in such a way that the device will have the default frequency set resistor connected from the RT/CLK pin to ground should the synchronization signal turn off. It is recommended to use a frequency set resistor connected as shown in Figure 42 through a 50Ω resistor to ground. The resistor should set the switching frequency close to the external CLK frequency. It is recommended to ac couple the synchronization signal through a 10 pF ceramic capacitor to RT/CLK pin and a 4kΩ series resistor. The series resistor reduces PH jitter in heavy load applications when synchronizing to an external clock and in applications which transition from synchronizing to RT mode. The first time the CLK is pulled above the CLKthresholdthedeviceswitchesfromtheRTresistorfrequencytoPLLmode.Theinternal0.5Vvoltagesource is removed and the CLK pin becomes high impedance as the PLL starts to lock onto the external signal. Since there is a PLL on the regulator the switching frequency can be higher or lower than the frequency set with the external resistor. The device transitions from the resistor mode to the PLL mode and then will increase or decreasetheswitchingfrequencyuntilthePLLlocksontotheCLKfrequencywithin100microseconds. When the device transitions from the PLL to resistor mode the switching frequency will slow down from the CLK frequency to 150 kHz, then reapply the 0.5V voltage and the resistor will then set the switching frequency. The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on VSENSE pin. The device implements a digital frequency shift to enable synchronizing to an external clock during normal startup and fault conditions. Figure 43, Figure 44 and Figure 45 show the device synchronized to an external system clockincontinuousconductionmode(ccm)discontinuousconduction(dcm)andpulseskipmode(psm). TPS54060A 10 pF 4 kW PLL R fset EXT RT/CLK Clock 50W Source Figure42. SynchronizingtoaSystemClock 22 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) PH PH EXT EXT IL IL Figure43.PlotofSynchronizinginCCM Figure44.PlotofSynchronizinginDCM PH EXT IL Figure45.PlotofSynchronizinginPSM 7.3.16 PowerGood(PWRGDPin) The PWRGD pin is an open drain output. Once the VSENSE pin is between 94% and 107% of the internal voltage reference the PWRGD pin is de-asserted and the pin floats. It is recommended to use a pull-up resistor between the values of 10 and 100kΩ to a voltage source that is 5.5V or less. The PWRGD is in a defined state once the VIN input voltage is greater than 1.5V but with reduced current sinking capability. The PWRGD will achievefullcurrentsinkingcapabilityasVINinputvoltageapproaches3V. ThePWRGDpinispulledlowwhentheVSENSEislowerthan92%orgreaterthan109%ofthenominalinternal reference voltage. Also, the PWRGD is pulled low, if the UVLO or thermal shutdown are asserted or the EN pin pulledlow. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 23 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) 7.3.17 OvervoltageTransientProtection The TPS54060A incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot when recovering from output fault conditions or strong unload transients on power supply designs with low value output capacitance. For example, when the power supply output is overloaded the error amplifier compares the actual output voltage to the internal reference voltage. If the VSENSE pin voltage is lower than the internal reference voltage for a considerable time, the output of the error amplifier will respond by clamping the error amplifier output to a high voltage. Thus, requesting the maximum output current. Once the condition is removed, the regulator output rises and the error amplifier output transitions to the steady state duty cycle. In some applications, the power supply output voltage can respond faster than the error amplifier output can respond, this actuality leads to the possibility of an output overshoot. The OVTP feature minimizes the output overshoot, when using a low value output capacitor, by implementing a circuit to compare the VSENSE pin voltage to OVTP threshold which is 109% of the internal voltage reference. If the VSENSE pin voltage is greater than the OVTP threshold, the high side MOSFET is disabled preventing current from flowing to the output and minimizing output overshoot. When the VSENSE voltage drops lower than the OVTP threshold, the high side MOSFET is allowed toturnonatthenextclockcycle. 7.3.18 ThermalShutdown The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 182°C. The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal trip threshold. Once the die temperature decreases below 182°C, the device reinitiates the power up sequence bydischargingtheSS/TRpin. 7.3.19 SmallSignalModelforLoopResponse Figure 46 shows an equivalent model for the TPS54060A control loop which can be modeled in a circuit simulation program to check frequency response and dynamic load response. The error amplifier is a transconductance amplifier with a gm of 97 μA/V. The error amplifier can be modeled using an ideal voltage EA controlled current source. The resistor R and capacitor C model the open loop gain and frequency response of o o the amplifier. The 1mV ac voltage source between the nodes a and b effectively breaks the control loop for the frequencyresponsemeasurements.Plottingc/ashowsthesmallsignalresponseofthefrequencycompensation. Plotting a/b shows the small signal response of the overall loop. The dynamic loop response can be checked by replacing R with a current source with the appropriate load step amplitude and step rate in a time domain L analysis.Thisequivalentmodelisonlyvalidforcontinuousconductionmodedesigns. PH V Power Stage O gm 1.9A/V ps a b R1 RESR COMP RL c VSENSE C 0.8 V OUT CO RO R3 gm ea C2 97mA/V R2 C1 Figure46. SmallSignalModelforLoopResponse 24 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) 7.3.20 SimpleSmallSignalModelforPeakCurrentModeControl Figure 47 describes a simple small signal model that can be used to understand how to design the frequency compensation. The TPS54060A power stage can be approximated to a voltage-controlled current source (duty cycle modulator) supplying current to the output capacitor and load resistor. The control to output transfer function is shown in Equation 14 and consists of a dc gain, one dominant pole, and one ESR zero. The quotient of the change in switch current and the change in COMP pin voltage (node c in Figure 46) is the power stage transconductance.Thegm fortheTPS54060Ais1.9A/V.Thelow-frequencygainofthepowerstagefrequency PS responseistheproductofthetransconductanceandtheloadresistanceasshowninEquation15. Astheloadcurrentincreasesanddecreases,thelow-frequencygaindecreasesandincreases,respectively.This variation with the load may seem problematic at first glance, but fortunately the dominant pole moves with the load current (see Equation 16). The combined effect is highlighted by the dashed line in the right half of Figure 47. As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover frequency the same for the varying load conditions which makes it easier to design the frequency compensation. The type of output capacitor chosen determines whether the ESR zero has a profound effect on the frequency compensation design. Using high ESR aluminum electrolytic capacitors may reduce the number frequency compensation components needed to stabilize the overall loop because the phase margin increases fromtheESRzeroatthelowerfrequencies(seeEquation17). VO VC Adc RESR fp RL gmps COUT fz Figure47.SimpleSmallSignalModelandFrequencyResponseforPeakCurrentModeControl æ s ö ç1+ ÷ VOUT = Adc´è 2p´fZ ø V æ s ö C ç1+ ÷ è 2p´fP ø (14) Adc=gm ´ R ps L (15) 1 f = P C ´R ´2p OUT L (16) 1 f = Z C ´R ´2p OUT ESR (17) 7.3.21 SmallSignalModelforFrequencyCompensation The TPS54060A uses a transconductance amplifier for the error amplifier and readily supports three of the commonly-used frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are shown in Figure 48. Type 2 circuits most likely implemented in high bandwidth power-supply designs using low ESR output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminum electrolytic or tantalum capacitors.. Equation 18 and Equation 19 show how to relate the frequency response of theamplifiertothesmallsignalmodelinFigure48.Theopen-loopgainandbandwidtharemodeledusingtheR O and C shown in Figure 48. See the application section for a design example using a Type 2A network with a O lowESRoutputcapacitor. Equation 18 through Equation 27 are provided as a reference for those who prefer to compensate using the preferred methods. Those who prefer to use prescribed method use the method outlined in the application sectionoruseswitchedinformation. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 25 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Feature Description (continued) V O R1 VSENSE gm Type 2A Type 2B Type 1 ea COMP Vref R3 C2 R3 R2 RO CO C2 C1 C1 Figure48. TypesofFrequencyCompensation Aol P1 A0 Z1 P2 A1 BW Figure49. FrequencyResponseoftheType2AandType2BFrequencyCompensation 26 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Feature Description (continued) Aol(V/V) Ro= gm ea (18) gm C = ea O 2p ´ BW (Hz) (19) æ s ö ç1+ ÷ è 2p´fZ1ø EA = A0´ æ s ö æ s ö ç1+ ÷´ç1+ ÷ è 2p´fP1ø è 2p´fP2 ø (20) R2 A0=gm ´ Ro ´ ea R1+R2 (21) R2 A1=gm ´ Ro||R3 ´ ea R1+R2 (22) 1 P1= 2p´Ro´C1 (23) 1 Z1= 2p´R3´C1 (24) 1 P2= type2a 2p ´ R3||R ´ (C2+C ) O O (25) 1 P2= type2b 2p ´ R3||R ´ C O O (26) 1 P2 = type 1 2p ´ R ´ (C2 + C ) O O (27) 7.4 Device Functional Modes 7.4.1 PulseSkipEco-Mode The TPS54060A operates in a pulse skip Eco mode at light load currents to improve efficiency by reducing switchingandgatedrivelosses.TheTPS54060Aisdesignedsothatiftheoutputvoltageiswithinregulationand the peak switch current at the end of any switching cycle is below the pulse skipping current threshold, the device enters Eco mode. This current threshold is the current level corresponding to a nominal COMP voltage or 500mV. When in Eco-Mode, the COMP pin voltage is clamped at 500mV and the high side MOSFET is inhibited. Further decreasesinloadcurrentorinoutputvoltagecannotdrivetheCOMPpinbelowthisclampvoltagelevel. Since the device is not switching, the output voltage begins to decay. As the voltage control loop compensates forthefallingoutputvoltage,theCOMPpinvoltagebeginstorise.Atthistime,thehighsideMOSFETisenabled andaswitchingpulseinitiatesonthenextswitchingcycle.ThepeakcurrentissetbytheCOMPpinvoltage.The output voltage re-charges the regulated value (see Figure 50), then the peak switch current starts to decrease, andeventuallyfallsbelowtheEcomodethresholdatwhichtimethedeviceagainentersEcomode. For Eco-mode operation, the TPS54060A senses peak current, not average or load current, so the load current where the device enters Eco mode is dependent on the output inductor value. For example, the circuit in Figure 51 enters Eco mode at about 20 mA of output current. When the load current is low and the output voltage is within regulation, the device enters a sleep mode and draws only 116μA input quiescent current. The internal PLL remains operating when in sleep mode. When operating at light load currents in the pulse skip mode,theswitchingtransitionsoccursynchronouslywiththeexternalclocksignal. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 27 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Device Functional Modes (continued) VOUT (ac) I L PH Figure50. PulseSkipModeOperation 7.4.2 NormalOperation When the input voltage is above the UVLO threshold and the EN voltage is above the enable threshold, the TPS54060A operates in normal switching modes. Normal continuous conduction mode (CCM) occurs when the minimum switch current is above 0 A or when the load current is above one half the peak to peak inductor ac ripplecurrent.Belowthatcurrent,theTPS54060Aoperatesindiscontinuousconductionmode(DCM). 7.4.3 StandbyOperation When the TPS54060A is operating in either normal CCM or Eco-Mode, they may be placed in standby by assertingtheENpinlow. 28 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validateandtesttheirdesignimplementationtoconfirmsystemfunctionality. 8.1 Application Information The TPS54060A device is a wide input voltage (3.5 V to 60 V), step-down, DC-DC converter with an adjustable output voltage and an output current of up to 0.5 A. Switching frequency is adjustable between 100 kHz to 2.5 MHzandhenceefficiencyandexternalcomponentsizecanbeoptimizedforeachapplication. 8.2 Typical Application The following design example shows a typical 3.3 V output application with an input voltage range of 12 V to 48 V.Themaximumoutputcurrentis0.5A. U1 TPS4060ADGQ VOUT 3.3 V, 0.5A VIN = 12 - 48 V C3 31.6k 73.2k 5.6pF 237k 3300pF Figure51. HighFrequency,3.3-VOutputPowerSupplyDesignwithAdjustedUVLO 8.2.1 DesignRequirements Forthisdesignexample,usetheparameterslistedinTable1astheinputparameters. Table1. DesignParameters DESIGNPARAMETERS EXAMPLEVALUE OutputVoltage 3.3V TransientResponse0to0.5Aloadstep ΔVout=4% MaximumOutputCurrent 1.5A InputVoltage 34Vnom.12Vto48V OutputVoltageRipple 1%ofVout StartInputVoltage(risingVIN) 8.9V StopInputVoltage(fallingVIN) 7.9V Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 29 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 8.2.2 DetailedDesignProcedure This example details the design of a high frequency switching regulator design using ceramic output capacitors. Afewparametersmustbeknowninordertostartthedesignprocess.Theseparametersaretypicallydetermined atthesystemlevel. 8.2.2.1 SelectingtheSwitchingFrequency The first step is to decide on a switching frequency for the regulator. Typically, the user will want to choose the highest switching frequency possible since this will produce the smallest solution size. The high switching frequency allows for lower valued inductors and smaller output capacitors compared to a power supply that switchesatalowerfrequency.Theswitchingfrequencythatcanbeselectedislimitedbytheminimumon-timeof theinternalpowerswitch,theinputvoltageandtheoutputvoltageandthefrequencyshiftlimitation. Equation 12 and Equation 13 must be used to find the maximum switching frequency for the regulator, choose the lower value of the two equations. Switching frequencies higher than these values will result in pulse skipping orthelackofovercurrentprotectionduringashortcircuit. The typical minimum on time, t , is 130 ns for the TPS54060A. For this example, the output voltage is 3.3 V onmin and the maximum input voltage is 48 V, which allows for a maximum switch frequency up to 616 kHz when including the inductor resistance, on resistance and diode voltage in Equation 12. To ensure overcurrent runaway is not a concern during short circuits in your design use Equation 13 or the solid curve in Figure 41 to determine the maximum switching frequency. With a maximum input voltage of 48 V, assuming a diode voltage of 0.5 V, inductor resistance of 130mΩ, switch resistance of 400mΩ, a current limit value of 0.94 A and a short circuitoutputvoltageof0.1V.Themaximumswitchingfrequencyisapproximately923kHz. Choosing the lower of the two values and adding some margin a switching frequency of 500kHz is used. To determinethetimingresistanceforagivenswitchingfrequency,useEquation11orthecurveinFigure39. TheswitchingfrequencyissetbyresistorR showninFigure51. 3 8.2.2.2 OutputInductorSelection(L ) O Tocalculatetheminimumvalueoftheoutputinductor,useEquation28. K isacoefficientthatrepresentstheamountofinductorripplecurrentrelativetothemaximumoutputcurrent. IND The inductor ripple current will be filtered by the output capacitor. Therefore, choosing high inductor ripple currents will impact the selection of the output capacitor since the output capacitor must have a ripple current rating equal to or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion ofthedesigner;however,thefollowingguidelinesmaybeused. For designs using low ESR output capacitors such as ceramics, a value as high as K = 0.3 may be used. IND When using higher ESR output capacitors, K = 0.2 yields better results. Since the inductor ripple current is IND part of the PWM control system, the inductor ripple current should always be greater than 30 mA for dependable operation. In a wide input voltage regulator, it is best to choose an inductor ripple current on the larger side. This allowstheinductortostillhaveameasurableripplecurrentwiththeinputvoltageatitsminimum. For this design example, use K = 0.3 and the minimum inductor value is calculated to be 39.7 μH. For this IND design, a nearest standard value was chosen: 47μH. For the output filter inductor, it is important that the RMS current and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from Equation30andEquation31. For this design, the RMS inductor current is 0.501 A and the peak inductor current is 0.563 A. The chosen inductorisaMSS1048-473ML.Ithasasaturationcurrentratingof1.44AandanRMScurrentratingof1.83A. As the equation set demonstrates, lower ripple currents will reduce the output voltage ripple of the regulator but will require a larger value of inductance. Selecting higher ripple currents will increase the output voltage ripple of theregulatorbutallowforalowerinductancevalue. The current flowing through the inductor is the inductor ripple current plus the output current. During power up, faults or transient load conditions, the inductor current can increase above the calculated peak inductor current level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of the device. For this reason, the most conservative approach is to specify an inductor with a saturation current ratingequaltoorgreaterthantheswitchcurrentlimitratherthanthepeakinductorcurrent. 30 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 Vinmax - Vout Vout Lomin= ´ Io ´ K Vinmax ´ ƒsw IND (28) IRIPPLE = VOUVTin´ma(xVin´mLaOx ´-fSVWOUT) (29) 2 I = (I )2 + 1 ´æçVOUT ´ (Vinmax - VOUT)ö÷ L(rms) O 12 ç Vinmax ´ L ´ f ÷ è O SW ø (30) Iripple ILpeak = Iout+ 2 (31) 8.2.2.3 OutputCapacitor There are three primary considerations for selecting the value of the output capacitor. The output capacitor will determine the modulator pole, the output voltage ripple, and how the regulators responds to a large change in loadcurrent.Theoutputcapacitanceneedstobeselectedbasedonthemorestringentofthesethreecriteria. The desired response to a large change in the load current is the first criteria. The output capacitor needs to supply the load with current when the regulator can not. This situation would occur if there are desired hold-up times for the regulator where the output capacitor must hold the output voltage above a certain level for a specified amount of time after the input power is removed. The regulator also will temporarily not be able to supply sufficient output current if there is a large, fast increase in the current needs of the load such as transitioning from no load to a full load. The regulator usually needs two or more clock cycles for the control loop to see the change in load current and output voltage and adjust the duty cycle to react to the change. The output capacitor must be sized to supply the extra current to the load until the control loop responds to the load change. The output capacitance must be large enough to supply the difference in current for 2 clock cycles while only allowing a tolerable amount of droop in the output voltage. Equation 32 shows the minimum output capacitance necessarytoaccomplishthis. Where ΔIout is the change in output current, ƒsw is the regulators switching frequency and ΔVout is the allowable change in the output voltage. For this example, the transient load response is specified as a 4% change in Vout for a load step from 0A (no load) to 0.5 A (full load). For this example, ΔIout = 0.5-0 = 0.5 A and ΔVout = 0.04 × 3.3 = 0.132 V. Using these numbers gives a minimum capacitance of 15.2μF. This value does not take the ESR of the output capacitor into account in the output voltage change. For ceramic capacitors, the ESR is usually small enough to ignore in this calculation. Aluminum electrolytic and tantalum capacitors have higherESRthatshouldbetakenintoaccount. The catch diode of the regulator can not sink current so any stored energy in the inductor will produce an output voltage overshoot when the load current rapidly decreases, see Figure 52. The output capacitor must also be sized to absorb energy stored in the inductor when transitioning from a high load current to a lower load current. The excess energy that gets stored in the output capacitor will increase the voltage on the capacitor. The capacitor must be sized to maintain the desired output voltage during these transient periods. Equation 33 is used to calculate the minimum capacitance to keep the output voltage overshoot to a desired value. Where L is the value of the inductor, I is the output current under heavy load, I is the output under light load, VF is the OH OL final peak output voltage, and Vi is the initial capacitor voltage. For this example, the worst case load step will be from 0.5 A to 0 A. The output voltage will increase during this load transition and the stated maximum in our specification is 4% of the output voltage. This will make Vf = 1.04 × 3.3 = 3.432. Vi is the initial capacitor voltage which is the nominal output voltage of 3.3 V. Using these numbers in Equation 33 yields a minimum capacitance of13.2μF. Equation 34 calculates the minimum output capacitance needed to meet the output voltage ripple specification. Where fsw is the switching frequency, V is the maximum allowable output voltage ripple, and I is the oripple ripple inductorripplecurrent.Equation34yields1μF. Equation 35 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple specification.Equation35indicatestheESRshouldbelessthan248mΩ. The most stringent criteria for the output capacitor is 15.2μF of capacitance to keep the output voltage in regulationduringanloadtransient. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 31 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com Additional capacitance de-ratings for aging, temperature and dc bias should be factored in which will increase thisminimumvalue.Forthisexample,a47 μF10VX5Rceramiccapacitorwith5mΩ ofESRwillbeused. Capacitors generally have limits to the amount of ripple current they can handle without failing or producing excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor datasheetsspecifytheRootMeanSquare(RMS)valueofthemaximumripplecurrent.Equation36 canbeused to calculate the RMS ripple current the output capacitor needs to support. For this application, Equation 36 yields 37.7mA. 2 ´ DIout Cout> ¦sw ´ DVout (32) (Ioh2 - Iol2) Cout>Lo ´ (V¦2 -Vi2) (33) 1 1 Cout> ´ 8 ´¦sw VORIPPLE I RIPPLE (34) V R < ORIPPLE ESR I RIPPLE (35) Vout ´ (Vinmax - Vout) Icorms= 12 ´ Vinmax ´ Lo ´ ¦sw (36) 8.2.2.4 CatchDiode The TPS54060A requires an external catch diode between the PH pin and GND. The selected diode must have a reverse voltage rating equal to or greater than V . The peak current rating of the diode must be greater IN(max) than the maximum inductor current. The diode should also have a low forward voltage. Schottky diodes are typically a good choice for the catch diode due to their low forward voltage. The lower the forward voltage of the diode,thehighertheefficiencyoftheregulator. Typically, the higher the voltage and current ratings the diode has, the higher the forward voltage will be. Since the design example has an input voltage up to 48V, a diode with a minimum of 60V reverse voltage will be selected. For the example design, the B160A Schottky diode is selected for its lower forward voltage and it comes in a largerpackagesizewhichhasgoodthermalcharacteristicsoversmalldevices.Thetypicalforwardvoltageofthe B160Ais0.50volts. The diode must also be selected with an appropriate power rating. The diode conducts the output current during the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by the forward voltage of the diode which equals the conduction losses of the diode. At higher switch frequencies, the ac losses of the diode need to be taken into account. The ac losses of the diode are due to the charging and discharging of the junction capacitance and reverse recovery. Equation 37 is used to calculate the total power dissipation,conductionlossesplusaclosses,ofthediode. The B160A has a junction capacitance of 110pF. Using Equation 37, the selected diode will dissipate 0.297 Watts. This power dissipation, depending on mounting techniques, should produce a 5.9°C temperature rise in thediodewhentheinputvoltageis48Vandtheloadcurrentis0.5A. If the power supply spends a significant amount of time at light load currents or in sleep mode consider using a diodewhichhasalowleakagecurrentandslightlyhigherforwardvoltagedrop. 2 (Vinmax - Vout) ´ Iout ´ Vƒd Cj ´ ƒsw ´ (Vin+ Vƒd) Pd= + Vinmax 2 (37) 32 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 8.2.2.5 InputCapacitor The TPS54060A requires a high quality ceramic, type X5R or X7R, input decoupling capacitor of at least 3 μF of effective capacitance and in some applications a bulk capacitance. The effective capacitance includes any dc bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The capacitormustalsohavearipplecurrentratinggreaterthanthemaximuminputcurrentrippleoftheTPS54060A. TheinputripplecurrentcanbecalculatedusingEquation38. The value of a ceramic capacitor varies significantly over temperature and the amount of dc bias applied to the capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors because they have a high capacitance to volume ratio and are fairly stable over temperature. The output capacitor must also be selected with the dc bias taken into account. The capacitance value of a capacitor decreasesasthedcbiasacrossacapacitorincreases. For this example design, a ceramic capacitor with at least a 60V voltage rating is required to support the maximum input voltage. Common standard ceramic capacitor voltage ratings include 4V, 6.3V, 10V, 16V, 25V, 50V or 100V so a 100V capacitor should be selected. For this example, two 2.2μF, 100V capacitors in parallel have been selected. Table 2 shows a selection of high voltage capacitors. The input capacitance value determinestheinputripplevoltageoftheregulator.TheinputvoltageripplecanbecalculatedusingEquation39. Using the design example values, Ioutmax = 0.5 A, Cin = 4.4μF, ƒsw = 500 kHz, yields an input voltage ripple of 57mVandarmsinputripplecurrentof0.223A. Vout (Vinmin - Vout) Icirms=Iout ´ ´ Vinmin Vinmin (38) Ioutmax ´ 0.25 ΔVin= Cin ´ ¦sw (39) Table2.CapacitorTypes VENDOR VALUE(μF) EIASize VOLTAGE DIALECTRIC COMMENTS 1.0to2.2 100V 1210 GRM32series 1.0to4.7 50V Murata 1.0 100V 1206 GRM31series 1.0to2.2 50V 1.0101.8 50V 2220 1.0to1.2 100V Vishay VJX7Rseries 1.0to3.9 50V 2225 1.0to1.8 100V X7R 1.0to2.2 100V 1812 CseriesC4532 1.5to6.8 50V TDK 1.0.to2.2 100V 1210 CseriesC3225 1.0to3.3 50V 1.0to4.7 50V 1210 1.0 100V AVX X7Rdielectricseries 1.0to4.7 50V 1812 1.0to2.2 100V 8.2.2.6 SlowStartCapacitor The slow start capacitor determines the minimum amount of time it will take for the output voltage to reach its nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This is also used if the output capacitance is large and would require large amounts of current to quickly charge the capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the TPS54060A reach the current limit or excessive current draw from the input power supply may cause the input voltagerailtosag.Limitingtheoutputvoltageslewratesolvesbothoftheseproblems. Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 33 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com The slow start time must be long enough to allow the regulator to charge the output capacitor up to the output voltage without drawing excessive current. Equation 40 can be used to find the minimum slow start time, tss, necessary to charge the output capacitor, Cout, from 10% to 90% of the output voltage, Vout, with an average slow start current of Issavg. In the example, to charge the 47μF output capacitor up to 3.3V while only allowing theaverageinputcurrenttobe0.125Awouldrequirea1msslowstarttime. Once the slow start time is known, the slow start capacitor value can be calculated using Equation 6. For the example circuit, the slow start time is not too critical since the output capacitor value is 47μF which does not require much current to charge to 3.3V. The example circuit has the slow start time set to an arbitrary value of 3.2mswhichrequiresa0.01μFcapacitor. Cout ´ Vout ´ 0.8 tss> Issavg (40) 8.2.2.7 BootstrapCapacitorSelection A 0.1-μF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. It is recommended to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have a 10V orhighervoltagerating. 8.2.2.8 UnderVoltageLockOutSetPoint The Under Voltage Lock Out (UVLO) can be adjusted using an external voltage divider on the EN pin of the TPS54060A. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power downorbrownoutswhentheinputvoltageisfalling.Fortheexampledesign,thesupplyshouldturnonandstart switching once the input voltage increases above 8.9V (enabled). After the regulator starts switching, it should continuetodosountiltheinputvoltagefallsbelow7.9V(UVLOstop). TheprogrammableUVLOandenablevoltagesaresetusingaresistordividerbetweenVinandgroundtotheEN pin. Equation 2 through Equation 3 can be used to calculate the resistance values necessary. For the example application,a332kΩ betweenVinandENanda56.2kΩ betweenENandgroundarerequiredtoproducethe8.9 and7.9voltstartandstopvoltages. 8.2.2.9 OutputVoltageandFeedbackResistorsSelection For the example design, 10.0 kΩ was selected for R2. Using Equation 1, R1 is calculated as 31.25 kΩ. The nearest standard 1% resistor is 31.6 kΩ. Due to current leakage of the VSENSE pin, the current flowing through the feedback network should be greater than 1 μA in order to maintain the output voltage accuracy. This requirement makes the maximum value of R2 equal to 800 kΩ. Choosing higher resistor values will decrease quiescentcurrentandimproveefficiencyatlowoutputcurrentsbutmayintroducenoiseimmunityproblems. 8.2.2.10 Compensation There are several methods used to compensate DC/DC regulators. The method presented here is easy to calculate and ignores the effects of the slope compensation that is internal to the device. Since the slope compensation is ignored, the actual cross over frequency will usually be lower than the cross over frequency used in the calculations. This method assume the crossover frequency is between the modulator pole and the esr zero and the esr zero is at least 10 times greater the modulator pole. Use SwitcherPro software for a more accuratedesign. To get started, the modulator pole, fpmod, and the esr zero, fz1 must be calculated using Equation 41 and Equation 42. For Cout, use a derated value of 40 μf. Use equations Equation 43 and Equation 44, to estimate a starting point for the crossover frequency, fco, to design the compensation. For the example design, fpmod is 603 Hz and fzmod is 796 kHz. Equation 43 is the geometric mean of the modulator pole and the esr zero and Equation 44 is the mean of modulator pole and the switching frequency. Equation 43 yields 21.9 kHz and Equation44gives12.3kHz.UsethelowervalueofEquation43 orEquation44foraninitialcrossoverfrequency. For this example, fco is 12.3kHz. Next, the compensation components are calculated. A resistor in series with a capacitor is used to create a compensating zero. A capacitor in parallel to these two components forms the compensatingpole. Ioutmax ¦pmod= 2×p ×Vout×Cout (41) 34 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 1 ¦zmod= 2 ´ p ´ Resr×Cout (42) f = f mod´ f mod co p z (43) f f = f mod´ sw co p 2 (44) To determine the compensation resistor, R4, use Equation 45. Assume the power stage transconductance, gmps, is 1.9A/V. The output voltage, Vo, reference voltage, VREF, and amplifier transconductance, gmea, are 3.3V, 0.8V and 97μA/V, respectively. R4 is calculated to be 72.6 kΩ, use the nearest standard value of 73.2kΩ. Use Equation 46 to set the compensation zero to the modulator pole frequency. Equation 46 yields 3600pF for compensatingcapacitorC7,a3300pFisusedontheboard. æ2´p´ f ´C ö æ V ö R4=ç co out ÷´ç out ÷ è gmps ø èVref´gmeaø (45) 1 C7= 2´p´R4´ f mod p (46) Use the larger value of Equation 47 and Equation 48 to calculate the C8, to set the compensation pole. Equation48yields8.7pFsotheneareststandardof10pFisused. C ´Resr C8= o R4 (47) 1 C8= R4´ f ´p sw (48) 8.2.2.11 PowerDissipationEstimate The following formulas show how to estimate the IC power dissipation under continuous conduction mode (CCM) operation.Theseequationsshouldnotbeusedifthedeviceisworkingindiscontinuousconductionmode(DCM). The power dissipation of the IC includes conduction loss (Pcon), switching loss (Psw), gate drive loss (Pgd) and supplycurrent(Pq). Vout Pcon=Io2 ´ R ´ DS(on) Vin (49) Psw=Vin2 ´ ¦sw ´ lo ´ 0.25 ´ 10-9 (50) Pgd=Vin ´ 3 ´ 10-9´¦sw (51) Pq=116 ´ 10-6 ´Vin where • Ioistheoutputcurrent(A). • R istheon-resistanceofthehigh-sideMOSFET(Ω). DS(on) • VOUTistheoutputvoltage(V). • VINistheinputvoltage(V). • fswistheswitchingfrequency(Hz). (52) So Ptot=Pcon+Psw +Pgd+Pq (53) ForgivenT , A TJ=TA+Rth ´ Ptot (54) ForgivenT =150°C JMAX TAmax=TJmax - Rth ´ Ptot where • Ptotisthetotaldevicepowerdissipation(W). Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 35 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com • T istheambienttemperature(°C). A • T isthejunctiontemperature(°C). J • Rthisthethermalresistanceofthepackage(°C/W). • T ismaximumjunctiontemperature(°C). JMAX • T ismaximumambienttemperature(°C). (55) AMAX Therewillbeadditionalpowerlossesintheregulatorcircuitduetotheinductoracanddclosses,thecatchdiode andtraceresistancethatwillimpacttheoverallefficiencyoftheregulator. 8.2.3 ApplicationCurves V = 20 V/div (AC Coupled) IN V = 1 V/div OUT I = 200 mA/div (0.125 to 0.375AStep) OUT V = 20 V/div IN Time = 5 ms/div Time = 2 ms/div Figure52.LoadTransient Figure53.StartupWithVIN V = 10 mV/div (AC Coupled) OUT V = 10 mV/div (AC Coupled) OUT PH = 20 V/div PH = 20 V/div Inductor Current = 100 mA/div Time = 1μs/div Time = 5μs/div Figure54.OutputRippleCCM Figure55.OutputRipple,DCM 36 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 V = 10 mV/div (AC Coupled) OUT V = 20 mV/div (AC Coupled) IN PH = 20 V/div PH = 20 V/div Time = 50μs/div Time = 1μs/div Figure56.OutputRipple,PSM Figure57.InputRippleCCM 100 V = 20 V/div (AC Coupled) IN 90 80 70 PH = 20 V/div ency (%) 5600 VIN= 34 VVIN= 24 VVIN= 18 VVIN= 12 V Effici 40 VIN= 42 V 30 Inductor Current = 100 mA/div 20 10 0 0 0.05 0.10 0.15 0.20 0.25 0.30 0.35 0.40 0.45 0.50 Time = 5μs/div Output Current (A) C028 Figure59.EfficiencyvsLoadCurrent Figure58.InputRippleDCM 100 60 90 150 40 80 100 70 Phase 20 %) 60 50 Efficiency ( 4500 VIN= 34 VVIN= 24 VVIN= 18 VVIN= 12 V Gain (dB) 0 Gain 0–50 oPhase () –20 30 V = 42 V IN –100 20 –40 –150 10 0 –60 0 0.02 0.04 0.06 0.08 0.10 100 1k 10k 100k 1M Output Current (A) Frequency (Hz) C029 C030 Figure60.LightLoadEfficiency Figure61.OverallLoopFrequencyResponse Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 37 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 0.1 0.1 0.08 0.08 0.06 0.06 0.04 0.04 Regulation (%) –00..00022 Regulation (%) –00..00022 –0.04 –0.04 –0.06 –0.06 –0.08 –0.08 –0.1 –0.1 0.00 0.25 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5 10 15 20 25 30 35 40 45 50 55 60 Load Current (A) Input Voltage (V) C031 C032 Figure62.RegulationvsLoadCurrent Figure63.RegulationvsInputVoltage 8.3 System Examples VIN + Cin Cboot Lo VIN BOOT PH GND Cd R1 GND + TPS54060A R2 Co VOUT VSENSE EN SS/TR COMP RT/CLK Rcomp Czero Cpole Css RT Figure64. InvertingPowerSupplyfromtheSLVA317 ApplicationNote 38 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 System Examples (continued) VOPOS + VIN Copos + Cin Cboot GND VIN BOOT PH Lo Cd R1 + GND Coneg TPS54060A R2 VONEG VSENSE EN SS/TR COMP RT/CLK Rcomp Czero Cpole Css RT Figure65. SplitRailPowerSupplyBasedontheSLVA369 ApplicationNote Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 39 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 9 Power Supply Recommendations TPS54060A is designed to operate from an input voltage supply range between 3.5 V and 60 V. This input supply should be well regulated. If the input supply is located more than a few inches from the TPS54060A converter additional bulk capacitance may be required in addition to the ceramic bypass capacitors. An electrolyticcapacitorwithavalueof100μFisatypicalchoice. 10 Layout 10.1 Layout Guidelines Layout is a critical portion of good power supply design. There are several signals paths that conduct fast changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise or degrade the power supplies performance. To help eliminate these problems, the VIN pin should be bypassed to ground with a low ESR ceramic bypass capacitor with X5R or X7R dielectric. Care should be taken to minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch diode. See Figure 66 for a PCB layout example. The GND pin should be tied directly to the power pad under the ICandthepowerpad. The power pad should be connected to any internal PCB ground planes using multiple vias directly under the IC. The PH pin should be routed to the cathode of the catch diode and to the output inductor. Since the PH connection is the switching node, the catch diode and output inductor should be located close to the PH pins, andtheareaofthePCBconductorminimizedtopreventexcessivecapacitivecoupling.Foroperationatfullrated load,thetopsidegroundareamustprovideadequateheatdissipatingarea.TheRT/CLKpinissensitivetonoise so the RT resistor should be located as close as possible to the IC and routed with minimal lengths of trace. The additional external components can be placed approximately as shown. It may be possible to obtain acceptable performance with alternate PCB layouts, however this layout has been shown to produce good results and is meantasaguideline. TheestimatedprintedcircuitboardareaforthecomponentsusedinthedesignofFigure51 is0.55in2.Thisarea doesnotincludetestpointsorconnectors. 40 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

TPS54060A www.ti.com SLVSB57C–MARCH2012–REVISEDJANUARY2016 10.2 Layout Example Vout Output Capacitor Output Topside Inductor Ground Route Boot Capacitor Catch Area Trace on another layer to Diode provide wide path for topside ground Input Bypass Capacitor BOOT PH Vin VIN GND EN COMP UVLO Adjust SS/TR VSENSE Compensation Resistor Resistors Network RT/CLK PWRGD Divider Slow Start Frequency Thermal VIA Capacitor Set Resistor Signal VIA Figure66. PCBLayoutExample Copyright©2012–2016,TexasInstrumentsIncorporated SubmitDocumentationFeedback 41 ProductFolderLinks:TPS54060A

TPS54060A SLVSB57C–MARCH2012–REVISEDJANUARY2016 www.ti.com 11 Device and Documentation Support 11.1 Device Support 11.1.1 Third-PartyProductsDisclaimer TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER ALONEORINCOMBINATIONWITHANYTIPRODUCTORSERVICE. 11.1.2 DevelopmentSupport FortheWEBENCHSoftwareTool,gotowww.TI.com/WEBENCH. 11.2 Community Resources The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TIE2E™OnlineCommunity TI'sEngineer-to-Engineer(E2E)Community.Createdtofostercollaboration amongengineers.Ate2e.ti.com,youcanaskquestions,shareknowledge,exploreideasandhelp solveproblemswithfellowengineers. DesignSupport TI'sDesignSupport QuicklyfindhelpfulE2Eforumsalongwithdesignsupporttoolsand contactinformationfortechnicalsupport. 11.3 Trademarks Eco-Mode,PowerPAD,SwitcherPro,E2EaretrademarksofTexasInstruments. WEBENCHisaregisteredtrademarkofTexasInstruments. Allothertrademarksarethepropertyoftheirrespectiveowners. 11.4 Electrostatic Discharge Caution Thesedeviceshavelimitedbuilt-inESDprotection.Theleadsshouldbeshortedtogetherorthedeviceplacedinconductivefoam duringstorageorhandlingtopreventelectrostaticdamagetotheMOSgates. 11.5 Glossary SLYZ022—TIGlossary. Thisglossarylistsandexplainsterms,acronyms,anddefinitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of thisdocument.Forbrowser-basedversionsofthisdatasheet,refertotheleft-handnavigation. 42 SubmitDocumentationFeedback Copyright©2012–2016,TexasInstrumentsIncorporated ProductFolderLinks:TPS54060A

PACKAGE OPTION ADDENDUM www.ti.com 3-Sep-2016 PACKAGING INFORMATION Orderable Device Status Package Type Package Pins Package Eco Plan Lead/Ball Finish MSL Peak Temp Op Temp (°C) Device Marking Samples (1) Drawing Qty (2) (6) (3) (4/5) TPS54060ADGQ ACTIVE MSOP- DGQ 10 80 Green (RoHS CU NIPDAUAG Level-1-260C-UNLIM -40 to 125 5406A PowerPAD & no Sb/Br) TPS54060ADGQR ACTIVE MSOP- DGQ 10 2500 Green (RoHS CU NIPDAUAG Level-1-260C-UNLIM -40 to 125 5406A PowerPAD & no Sb/Br) TPS54060ADRCR ACTIVE VSON DRC 10 3000 Green (RoHS CU NIPDAU Level-1-260C-UNLIM -40 to 125 5406A & no Sb/Br) TPS54060ADRCT ACTIVE VSON DRC 10 250 Green (RoHS CU NIPDAU Level-1-260C-UNLIM -40 to 125 5406A & no Sb/Br) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Addendum-Page 1

PACKAGE OPTION ADDENDUM www.ti.com 3-Sep-2016 Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 2

PACKAGE MATERIALS INFORMATION www.ti.com 3-Aug-2017 TAPE AND REEL INFORMATION *Alldimensionsarenominal Device Package Package Pins SPQ Reel Reel A0 B0 K0 P1 W Pin1 Type Drawing Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant (mm) W1(mm) TPS54060ADGQR MSOP- DGQ 10 2500 330.0 12.4 5.3 3.3 1.3 8.0 12.0 Q1 Power PAD TPS54060ADRCR VSON DRC 10 3000 330.0 12.4 3.3 3.3 1.0 8.0 12.0 Q2 TPS54060ADRCT VSON DRC 10 250 180.0 12.4 3.3 3.3 1.0 8.0 12.0 Q2 PackMaterials-Page1

PACKAGE MATERIALS INFORMATION www.ti.com 3-Aug-2017 *Alldimensionsarenominal Device PackageType PackageDrawing Pins SPQ Length(mm) Width(mm) Height(mm) TPS54060ADGQR MSOP-PowerPAD DGQ 10 2500 346.0 346.0 35.0 TPS54060ADRCR VSON DRC 10 3000 346.0 346.0 35.0 TPS54060ADRCT VSON DRC 10 250 203.0 203.0 35.0 PackMaterials-Page2

None

None

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GENERIC PACKAGE VIEW DRC 10 VSON - 1 mm max height PLASTIC SMALL OUTLINE - NO LEAD Images above are just a representation of the package family, actual package may vary. Refer to the product data sheet for package details. 4204102-3/M

PACKAGE OUTLINE DRC0010J VSON - 1 mm max height SCALE 4.000 PLASTIC SMALL OUTLINE - NO LEAD 3.1 B A 2.9 PIN 1 INDEX AREA 3.1 2.9 1.0 C 0.8 SEATING PLANE 0.05 0.00 0.08 C 1.65 0.1 2X (0.5) (0.2) TYP EXPOSED 4X (0.25) THERMAL PAD 5 6 2X 11 SYMM 2 2.4 0.1 10 1 8X 0.5 0.30 10X 0.18 PIN 1 ID SYMM 0.1 C A B (OPTIONAL) 0.5 0.05 C 10X 0.3 4218878/B 07/2018 NOTES: 1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing per ASME Y14.5M. 2. This drawing is subject to change without notice. 3. The package thermal pad must be soldered to the printed circuit board for optimal thermal and mechanical performance. www.ti.com

EXAMPLE BOARD LAYOUT DRC0010J VSON - 1 mm max height PLASTIC SMALL OUTLINE - NO LEAD (1.65) (0.5) 10X (0.6) 1 10 10X (0.24) 11 SYMM (2.4) (3.4) (0.95) 8X (0.5) 6 5 (R0.05) TYP ( 0.2) VIA TYP (0.25) (0.575) SYMM (2.8) LAND PATTERN EXAMPLE EXPOSED METAL SHOWN SCALE:20X 0.07 MIN 0.07 MAX EXPOSED METAL ALL AROUND ALL AROUND EXPOSED METAL SOLDER MASK METAL METAL UNDER SOLDER MASK OPENING SOLDER MASK OPENING NON SOLDER MASK SOLDER MASK DEFINED DEFINED (PREFERRED) SOLDER MASK DETAILS 4218878/B 07/2018 NOTES: (continued) 4. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature number SLUA271 (www.ti.com/lit/slua271). 5. Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown on this view. It is recommended that vias under paste be filled, plugged or tented. www.ti.com

EXAMPLE STENCIL DESIGN DRC0010J VSON - 1 mm max height PLASTIC SMALL OUTLINE - NO LEAD 2X (1.5) (0.5) SYMM EXPOSED METAL 11 TYP 10X (0.6) 1 10 (1.53) 10X (0.24) 2X (1.06) SYMM (0.63) 8X (0.5) 6 5 (R0.05) TYP 4X (0.34) 4X (0.25) (2.8) SOLDER PASTE EXAMPLE BASED ON 0.125 mm THICK STENCIL EXPOSED PAD 11: 80% PRINTED SOLDER COVERAGE BY AREA SCALE:25X 4218878/B 07/2018 NOTES: (continued) 6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate design recommendations. www.ti.com

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