MIC3223 High Power Boost LED Driver with Integrated FET General Description Features The MIC3223 is a constant current boost LED driver • 4.5V to 20V supply voltage capable of driving a series string of high power LEDs. The • 200mV feedback voltage with an accuracy of ±5% MIC3223 can be used in general lighting, bulb replacement, • Step-up output voltage (boost) conversion up to 37V garden pathway lighting and other solid state illumination • 1MHz switching frequency applications. • 100mΩ/3.5A internal power FET switch The MIC3223 is a peak current mode control PWM boost • LEDs can be dimmed using a PWM signal regulator and the 4.5V and 20V operating input voltage • User settable LED current (through external resistor) range allows multiple applications from a 5V or a 12V bus. • Externally programmable soft-start The MIC3223 implements a fixed internal 1MHz switching • Protection features that include: frequency to allow for a reduction in the design footprint size. Power consumption has been minimized through the – Output over-voltage protection (OVP) implementation of a 200mV feedback voltage that provides – Under-voltage lockout (UVLO) an accuracy of ±5%. The MIC3223 can be dimmed through – Over temperature protection the use of a PWM signal and features an enable pin for a • Junction temperature range: -40°C to +125°C low power shutdown state. • Available in a exposed pad 16-pin TSSOP package The MIC3223 is a very robust LED driver and offers the following protection features: over voltage protection (OVP), Applications thermal shutdown, switch current limiting and under voltage lockout (UVLO). • Architectural lighting The MIC3223 is offered in a low profile exposed pad 16-pin • Industrial lighting TSSOP package. • Signage Data sheets and support documentation can be found on • Landscape lighting (garden/pathway) Micrel’s web site at: www.micrel.com. • Under cabinet lighting • MR-16 bulbs _______________________________________________________________________________________________________________________ Typical Application Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com January 2010 M9999-011510-A

Micrel, Inc. MIC3223 Ordering Information Part Number Junction Temp. Range Package Lead Finish MIC3223YTSE –40° to +125°C 16-pin ePad TSSOP PB- free Pin Configuration 16-Pin ePad TSSOP (TSE) Pin Description Pin Number Pin Name Pin Function 1 EN Enable (Input): Logic high enables and logic low disables operation. 2 SS Soft Start (Input resistance of 30k). Connect a capacitor to GND for soft-start. Clamp the pin to a known voltage to control the internal reference voltage and hence the output current. 3 COMP Compensation Pin (Input): Add external R and C-to-GND to stabilize the converter. 4 FB Negative Input to Error Amp 5 OVP Connect to the centre tap of an external resistor divider, the top of which is tied to Vout and bottom-to-ground. 6 PGND Power Ground 7,8,9,10 SW Switch Node (Input): Internal NMOS switch Drain Pin 11 VIN Input Supply 12 DRVVDD For 4.5V < VIN < 6V, connect DRVVDD to VIN. DRVVDD is the input voltage supply for the converter’s internal power FET gate driver. For VIN > 6V, connect this pin to VDD. 13 VDD For 4.5V < VIN < 6V, this pin becomes the input voltage supply for the converter’s internal circuit. For VIN > 6V, this pin is an output of the internal 5.5V regulator that supplies internal circuits. User must add 10µF decoupling capacitor from VDD-to-AGND. 14 DIM_IN PWM input to control LED dimming. 15 DIM_OUT Output driver to drive external FET for LED dimming. 16 AGND Analog Ground 17 EP Connect to Power Ground January 2010 2 M9999-011510-A

Micrel, Inc. MIC3223 Absolute Maximum Ratings(1) Operating Ratings(2) Supply Voltage (V ).....................................................+22V Supply Voltage (V )......................................+4.5V to +20V IN IN Switch Voltage (V )..................................... -0.3V to +42V Switch Voltage (V )....................................................+37V SW SW Regulated Voltage (V )...............................-0.3V to +6.5V Junction Temperature (T ).........................-40°C to +125°C DD J Dimming In Voltage (V )...............-0.3V to (V + 0.3V) Junction Thermal Resistance DIM_IN DD Dimming Out Voltage (V )..........-0.3V to (V + 0.3V) ePad TSSOP-16L (θ )...................................36.5°C/W DIM_OUT DD JA Soft-Start Voltage (V ).......................-0.3V to (V + 0.3V) SS DD Enable Voltage (V )............................-0.3V to (V + 0.3V) EN IN Feedback Voltage (V )......................-0.3V to (V + 0.3V) FB DD Switch Current (I )..................................Internally Limited SW Comp Voltage (V ).......................-0.3V to (+V + 0.3V) COMP DD FET Driver Supply (V ).........................-0.3V to +6.5V DRVVDD PGND to AGND............................................-0.3V to +0.3V Over Voltage Protection (V )...........-0.3V to (V + 0.3V) OVP DD Peak Reflow Temperature (soldering, 10-20sec.).....260°C Storage Temperature (T )..........................-65°C to +150°C S ESD Rating(3)................................................................+2kV Electrical Characteristics(4) V = V = 12V; L = 22µH, C =4.7µF, C =2x4.7µF; T = 25°C, BOLD values indicate –40°C≤ T ≤ +125°C, unless otherwise noted. IN EN IN OUT A J Symbol Parameter Condition Min Typ Max Units V Voltage Supply Range 4.5 20 V IN V Under Voltage Lockout Monitoring for V 3 3.7 4.4 V UVLO DD V Over Voltage Protection 1.216 1.28 1.344 V OVP I Quiescent Current V =250mV 2.1 5 mA VIN FB I Shutdown Current V =0V 10 µA SD EN Room Temperature 190 200 210 mV V Feedback Voltage FB Over Temperature 184 216 mV I Feedback Input Current V =200mV -450 nA FB FB V Internal Voltage Regulator 5.3 V DD D Maximum Duty Cycle 85 90 95 % MAX V Line Regulation V =18V, V =8V to 16V, I =350mA 0.5 % DD LED IN LED I Switch Current Limit 3.5 9 10.5 A SW R Switch R plus R 100 mΩ SW DSON CS I Switch Leakage Current V =0, V =37V 0.01 10 µA SW EN SW Turn On 1.5 V V Enable Threshold EN Turn Off 0.4 V I Enable Pin Current 20 40 µA EN V DIM_IN Threshold High Logic High 1.5 V DIM_TH_H V DIM_IN Threshold Low Logic Low 0.4 V DIM_TH_L Hys DIM_IN Hysteresis 500 mV I DIM_IN Pin Current V = 5V 1 µA DIM_IN DIM_IN T Dim Delay (Rising) DIM_IN Rising 40 ns DR T Dim Delay (Falling DIM_IN Falling 30 ns DF January 2010 3 M9999-011510-A

Micrel, Inc. MIC3223 Symbol Parameter Condition Min Typ Max Units DIM_IN =1µs C = 1.25nF 0.7 1.3 µs DIM_OUT DIM MIN Minimum Dimming Pulse DIM_OUT measured from 4V rising to 2.5 0.5 1.5 µs falling DIM_OUT pull up resistance R DIM_OUT Resistance High 70 Ω DO I = +2mA DIM_OUT Dim Out pull down resistance R DIM_OUT Resistance Low 40 Ω DO I = -2mA DIM_OUT F Oscillator Frequency 0.7 1 1.3 MHz SW R Soft Start Resistance 30 46 62 kΩ SS Over Temperature Threshold Temperature rising 165 °C T SD Shutdown Hysteresis 10 °C Notes 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF. 4. Specification for packaged product only. Test Circuit January 2010 4 M9999-011510-A

Micrel, Inc. MIC3223 Typical Characteristics Efficiency VDD Voltage Current Limit vs. Input Voltage vs. Input Voltage vs. Input Voltage 98 5.50 9.5 96 T = 25°C 5.45 94 5.40 T = 25°C 9.0 EFFICIENCY (%)888899246802 VIOOUUTT = = 0 2.55AV VDD VOLTAGE (V)5555555.......01122335050505 VIOOUUTT = = 0 2.55AV CURRENT LIMIT (A)788...505 T = 25°CVIN = 4.5V to 6V 80 5.00 7.0 5 10 15 20 5 10 15 20 4 9 14 19 INPUT VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V) Feedback Voltage Switching Frequency Feedback Voltage vs. Input Voltage vs. Input Voltage vs. Temperature 0.210 1.2 0.220 0.208 0.218 REFERENCE VOTLAGE (V)00000000........111122229999000024680246 VIOOUUTT = =0 .3306VA SWITCHING FREQUENCY (MHz)1111....0011 T = 25°CVVVIODIOUNDUT =T == = 4 V0. 53I.N3V06V tAo 6V FEEDBACK VOLTAGE (V)00000000........222222220000111124680246 VVIOIOUNUT =T = =1 02 2.V366VA 0.190 0.9 0.200 4 9 14 19 4 9 14 19 -40 -20 0 20 40 60 80 100 120 INPUT VOLTAGE (V) INPUT VOLTAGE (V) TEMPERATURE (°C) Current Limit RSW_NODE vs. Switching Frequency vs. Temperature Temperature vs. Temperature 11.0 0.18 1.20 10.5 0.17 1.15 CURRENT LIMIT (A)106778899........05050505 VIN = 12V RSW_NODE (Ω)000000......111111123456 VVISIOWNU ==T =112 .33V6AV SWITCHING FREQUENCY (MHz)000111......899001505050 VVIOIOUNUT =T = =1 02 2.V366VA 6.0 0.10 0.80 -40 -20 0 20 40 60 80 100 120 -40 -20 0 20 40 60 80 100 120 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) Efficiency Efficiency Efficiency vs. Output Current vs. Output Current vs. Output Current 96 96 98 94 12V 94 16V 96 20V 92 92 14V 94 18V NCY (%)8980 10V 8V NCY (%)8980 NCY (%)9902 EFFICIE8846 EFFICIE8846 EFFICIE888468 82 VOUT = 25V 82 VOUT = 25V 82 VOUT = 25V 80 80 80 0 0.5 1 1.5 0 0.5 1 1.5 0 0.5 1 1.5 OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A) January 2010 5 M9999-011510-A

Micrel, Inc. MIC3223 Typical Characteristics (continued) Efficiency Efficiency vs. Output Current vs. Output Current 96 96 94 94 92 92 EFFICIENCY (%)88894680 10V EFFICIENCY (%)88894680 12V 82 VOUT = 25V 82 VOUT = 25V 80 80 0 0.5 1 1.5 0 0.5 1 1.5 OUTPUT CURRENT (A) OUTPUT CURRENT (A) January 2010 6 M9999-011510-A

Micrel, Inc. MIC3223 Functional Characteristics January 2010 7 M9999-011510-A

Micrel, Inc. MIC3223 Functional Characteristics (continued) January 2010 8 M9999-011510-A

Micrel, Inc. MIC3223 Functional Diagram January 2010 9 M9999-011510-A

Micrel, Inc. MIC3223 Functional Description current is regulated. If V drops, V increases and FB EA therefore the power FET remains on longer so that V CS A constant current output converter is the preferred can increase to the level of V . The reverse occurs EA method for driving LEDs. Small variations in current when V increases. FB have a minimal effect on the light output, whereas small variations in voltage have a significant impact on light PWM Dimming output. The MIC3223 LED driver is specifically designed This control process just described occurs during each to operate as a constant current LED Driver. DIM_IN pulse and when ever DIM_IN is high. When The MIC3223 is designed to operate as a boost DIM_IN is low, the boost converter will no longer switch converter, where the output voltage is greater than the and the output voltage will drop. For high dimming ratios input voltage. This configuration allows for the design of use an external PWM Dimming switch as shown in the driving multiple LEDs in series to help maintain color and Typical Application. When the dim pulse is on the brightness. The MIC3223 can also be configured as a external switch is on and circuit operates in the closed SEPIC converter, where the output voltage can be either loop control mode as described. When the DIM_IN is low above or below the input voltage. the boost converter does not switch and the external The MIC3223 has an input voltage range, from 4.5V and switch is open and no LED current can flow and the 20V, to address a diverse range of applications. In output voltage does not droop. When DIM_IN goes high addition, the LED current can be programmed to a wide the external switch is driven on and LED current flows. range of values through the use of an external resistor. The output voltage remains the same (about the same) This provides design flexibility in adjusting the current for during each on and off DIM_IN pulse. a particular application need. PWM Dimming can also be used in the Test Circuit in The MIC3223 features a low impedance gate driver applications that do not require high dimming ratios. In capable of switching large MOSFETs. This low the Test Circuit, the load is not removed from the output impedance provides higher operating efficiency. voltage between DIM_IN pulses and will therefore drain the output capacitors. The voltage that the output will The MIC3223 can control the brightness of the LEDs via discharge to is determined by the sum of the V (forward its PWM dimming capability. Applying a PWM signal (up F voltage drops of the LEDs). When V can no longer to 20kHz) to the DIM_IN pin allows for control of the OUT forward bias the LEDs, then the LED current will stop brightness of the LEDs. and the output capacitors will stop discharging. During The MIC3223 boost converter employs peak current the next DIM_IN pulse V has to charge back up OUT mode control. Peak current mode control offers before the full LED current will flow. For applications that advantages over voltage mode control in the following do not require high dimming ratios. manner. Current mode control can achieve a superior line transient performance compared to voltage mode control and is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. Page 9 of this datasheet shows the functional block diagram. Boost Converter operation The boost converter is a peak current mode pulse width modulation (PWM) converter and operates as follows. A flip-flop (FF) is set on the leading edge of the clock cycle. When the FF is set, a gate driver drives the power FET on. Current flows from V through the inductor (L) IN and through the power switch and also through the current sense resistor to PGND. The voltage across the current sense resistor is added to a slope compensation ramp (needed for stability). The sum of the current sense voltage and the slope compensation voltages (called V ) is fed into the positive terminal of the PWM CS comparator. The other input to the PWM comparator is the error amp output (called V ). The error amp’s EA negative input is the feedback voltage (V ). V is the FB FB voltage across R (R5). In this way the output LED ADJ January 2010 10 M9999-011510-A

Micrel, Inc. MIC3223 Application Information Output Over Voltage Protection (OVP) The MIC3223 provides an OVP circuitry in order to Constant Output Current Converter protect the system from an overvoltage fault condition. This OVP threshold can be programmed through the The MIC3223 is a peak current mode boost converter designed to drive high power LEDs with a constant use of external resistors (R3 and R4 in the Typical current output. The MIC3223 operates with an input Application). A reference value of 1.245V is used for the OVP. Equation 3 can be used to calculate the resistor voltage range from 4.5V to 20V. In the boost configuration, the output can be set from V up to 37V. value for R9 to set the OVP point. Normally use 100k for IN R3. The peak current mode control architecture of the MIC3223 provides the advantages of superior line R3 Eq. (3) R4= transient response as well as an easier to design (V /1.245)−1 OVP compensation. VDD The MIC3223 LED driver features a built-in soft start An internal linear regulator is used to provide the circuitry in order to prevent start-up surges. Other necessary internal bias voltages. When V is 6V or protection features include: IN below connect the V pin to V . Use a 10µF ceramic DD IN • Current Limit (I ) – Current sensing for over LIMIT bypass capacitor. current and overload protection • Over Voltage Protection (OVP) – output over DRVVDD voltage protection to prevent operation above a An internal linear regulator is used to provide the safe upper limit necessary internal bias voltages to the gate driver that drives the external FET. When V is above 6V connect • Under Voltage Lockout (UVLO) – UVLO designed IN DRVVDD to VDD. to prevent operation below a safe lower limit When V is 6V or below connect the DRVVDD pin to IN Setting the LED Current VIN. Use a bypass capacitor, 10µF ceramic capacitor. The current through the LED string is set via the value UVLO chosen for the current sense resistor R which is R5 in ADJ the schematic of the Typical Application. This value can Internal under voltage lock out (UVLO) prevents the part be calculated using Equation 1: from being used below a safe V voltage. The UVLO is IN 3.7V. Operation below 4.5V is not recommended. 0.2V Eq. (1) ILED= R Soft Start ADJ Another important parameter to be aware of in the boost Soft start is employed to lessen the inrush currents converter design is the ripple current. The amount of during turn on. At turn on the following occurs; ripple current through the LED string is equal to the 1. After about 1.5ms C will start to rise in a SS output ripple voltage divided by the LED AC resistance exponential manner according to; (R – provided by the LED manufacturer) plus the LED ⎛ −t ⎞ criuprprleen tt hsreonusgeh rtehsei sLtoErD R sADtrJi.n gT hies daempoeunndte notf uaplloown atbhlee VSS =0.2⎜⎜1−e(37kΩ×CSS)⎟⎟ ⎜ ⎟ application and is left to the designer’s discretion. The ⎝ ⎠ equation is shown in Equation 2. 2. According to the block diagram, V is the ref SS V node of the error amp. PWM switching start OUT Eq. (2) ΔI ≈ RIPPLE LED (R +R ) when VSS begins to rise. LED ADJ 3. When the C is fully charged, 0.2V will be at the SS I ×D Where V = LED error amp reference and steady state operation OUTRIPPLE C ×F begins. OUT SW 4. Design for soft-start time using the above Reference Voltage equation. The voltage feedback loop the MIC3223 uses an internal voltage of 200mV with an accuracy of ±5%. The feedback voltage is the voltage drop across the current sense resistor as shown in the Typical Application. When in regulation the voltage at V will equal 200mV. FB January 2010 11 M9999-011510-A

Micrel, Inc. MIC3223 If high dimming ratios are required, a lower Dimming frequency is required. During each DIM_IN pulse the inductor current has to ramp up to it steady state value in order for the programmed LED current to flow. The smaller the inductance value the faster this time is and a narrower DIM_IN pulse can be achieved. But smaller inductance means higher ripple current. Figure 1. Soft start LED Dimming The MIC3223 LED driver can control the brightness of the LED string via the use of pulse width modulated (PWM) dimming. An input signal from DC up to 20kHz can be applied to the DIM_IN pin (see Typical Figure 3. PWM Dimming 20% Application) to pulse the LED string ON and OFF. It is recommended to use PWM dimming signals above Figure 3 shows that switching occurs only during DIM_IN 120Hz to avoid any recognizable flicker by the human eye. PWM dimming is the preferred way to dim an LED on pulses. When DIM_IN is low the boost converter stops switching and the external LED is turned off. The in order to prevent color/wavelength shifting. Color LED current flows only when DIM_IN is high. Figure 3 wavelength shifting will occur with analog dimming. By employing PWM Dimming the output current level shows that the compensation pin (VCOMP) does not discharge between DIM_IN pulses. Therefore, when the remains constant during each DIM_IN pulse. The boost DIM_IN pulse starts again the converter resumes converter switches only when DIM_IN is high. Between DIM_IN pulses the output capacitors will slowly operation at the same VCOMP voltage. This eliminates the need for the comp pin to charge up during each DIM_IN discharge. The higher the DIM_IN frequency the less the pulse and allows for high Dimming ratios. output capacitors will discharge. PWM Dimming Limits The minimum pulse width of the DIM_IN is determined by the DIM_IN frequency and the L and C used in the boost stage output filter. At low DIM_IN frequencies lower dimming ratios can be achieved. LED_ON_TIME Dim_ratio = PERIOD PWMD Figure 4. PWM Dimming 10% and I 100Hz LED Figure 2. DIM_IN Dimming Ratio January 2010 12 M9999-011510-A

Micrel, Inc. MIC3223 Figure 5. PWM Dimming 20% and I 1kHz Figure 7. 5µs DIM_IN Pulse LED In Figure 4 is at 100Hz dimming frequency and Figure 5 Figure 7 shows the minimum DIM_IN pulse at these is 1kHz dimming frequency. The time it takes for the operating conditions before the I current starts to drop LED LED current to reach it full value is longer with a lower due to low V . The converter is ON (switching) only OUT Dimming frequency. The reason is the output capacitors during a DIM_IN pulse. slowly discharge between dimming pulses. Figure 7 shows that at this DIM_IN pulse width the converter is ON (switching) long enough to generate the necessary V to forward bias the LED string at the OUT programmed current level. Therefore this condition will result in the desired I . LED Figure 6. PWM Dimming 20% and I 1kHz LED Figure 6 shows the output voltage V discharge OUT between DIM_IN pulses. The amount of discharge is dependent on the time between DIM_IN pulses. Figure 8. 2.5µs DIM_IN Pulse Figure 8 shows that at this DIM_IN pulse width the converter in not ON (switching) long enough to generate the necessary V to forward bias the LED string at the OUT programmed current level. As a result the LED current drops. Therefore, this condition will not result in the desired I . LED January 2010 13 M9999-011510-A

Micrel, Inc. MIC3223 Design Procedure for a LED Driver Symbol Parameter Min Nom Max Units Input V Input Voltage 8 12 14 V IN I Input Current 2 A IN Output LEDs Number of LEDs 5 6 7 V Forward Voltage of LED 3.2 3.5 4.0 V F V Output Voltage 16 21 28 V OUT I LED Current 0.33 0.35 0.37 A LED I Required I Ripple 40 mA PP Pout Output Power 10.36 W DIM_IN PWM Dimming 0 100 % OVP Output Over Voltage Protection 30 V System F Switching Frequency 1 MHz SW eff Efficiency 80 % V Forward drop of schottky diode 0.5 V DIODE Table 1. Design example parameters January 2010 14 M9999-011510-A

Micrel, Inc. MIC3223 Design Example Using Equation 5, the following values have been calculated: In this example, we will be designing a boost LED driver operating off a 12V input. This design has been created to V ×I drive 6 LEDs at 350mA with a ripple of about 20%. We are IIN_RMS(max) = OUeT(fmfa×x)V OUT(max) =1.54A(RMS) designing for 80% efficiency at a switching frequency of IN(min) 1MHz. V ×I Eq (5) I = OUT(nom) OUT(nom) =0.74A Select RADJ IN_RMS(nom) eff×VIN(nom) (RMS) Having chosen the LED drive current to be 350mA in this V ×I example, the current can be set by choosing the RADJ I = OUT(min) OUT(min) =0.46A resistor from Equation 1: IN_RMS(min) eff×V (RMS) IN(max) 0.2V I is the same as I . R = =0.57Ω OUT LED ADJ 0.35A Selecting the inductor current (peak-to-peak), I , to be L_PP Use the next lowest standard value 0.56Ω. between 20% to 50% of I , in this case 40%, we IN_RMS(nom) obtain: I = 0.36A LED I = 0.4 × I = 0.4 × 0.74 = 0.30A The power dissipation in this resistor is: IN_PP(nom) IN_RMS(nom) P-P It can be difficult to find large inductor values with high P =ILED2 ×R =71mW RADJ ADJ saturation currents in a surface mount package. Due to Use a resistor rated at quarter watt or higher. this, the percentage of the ripple current may be limited by the available inductor. It is recommended to operate in the Operating Duty Cycle continuous conduction mode. The selection of L described The operating duty cycle can be calculated using Equation here is for continuous conduction mode. four provided below: V ×D ( ) Eq. (6) L= IN V −V +V Eq. (4) D= OUT IN DIODE IIN_PP×FSW V +V OUT DIODE Using the nominal values, we get: V is the V of the output diode D1 in the Typical DIODE f Application. It is recommended to use a schottky diode 12V×0.44 L= =18μH because it has a lower V than a junction diode. f 0.3A×1MHz These can be calculated for the nominal (typical) operating Select the next higher standard inductor value of 22µH. conditions, but should also be understood for the minimum Going back and calculating the actual ripple current gives: and maximum system conditions as listed below. (V −V +V ) VIN(min) ×Dmax 8V×0.72 Dnom = OUT(nom) IN(nom) DIODE IIN_PP(max) = = =0.26APP L×F 22μH×1MHz V +V SW OUT(nom) DIODE ( ) The average input current is different than the RMS input V −V +V Dmax = OUT(max) IN(min) DIODE current because of the ripple current. If the ripple current is VOUT(max) +VDIODE low, then the average input current nearly equals the RMS ( ) input current. In the case where the average input current V −V +V Dmin = OUT(min) IN(max) DIODE is different than the RMS, equation 7 shows the following: V +V OUT(min) DIODE (I )2 (21−12−0.5) Eq. (7) I = (I )2 − IN_PP Dnom = =0.44 IN_AVE(max) IN_RMS(max) 12 21+0.5 (21-12+0.5) I = (1.54)2−(0.24)2 ≈1.54A Dnom = =0.44 IN_AVE(max) 12 21+0.5 The Maximum Peak input current I can found using L_PK Therefore D = 44%, D = 72% and D = 15%. nom max min Equation 8: Inductor Selection Eq. (8) I = I + 0.5 ×I = 1.67A L_PK(max) IN_AVE(max) L_PP(max) First calculate the RMS input current (nominal, min and The saturation current (I ) at the highest operating SAT max) for the system given the operating conditions listed in temperature of the inductor must be rated higher than this. the design example table. The minimum value of the RMS The power dissipated in the inductor is: input current is necessary to ensure proper operation. January 2010 15 M9999-011510-A

Micrel, Inc. MIC3223 Eq. (9) PINDUCTOR = IIN_RMS(max)2 × DCR IIN_PP (0.3A) C = = =0.75μF A Coilcraft # MSS1260-223ML is used in this example. Its IN V ×F 8×50mV×1MHz IN(ripple) SW DCR is 52mΩ, I =2.7A SAT P = 1.542 × 52 mΩ = 0.123W This is the minimum value that should be used. To protect INDUCTOR the IC from inductive spikes or any overshoot, a larger Output Capacitor value of input capacitance may be required. In this LED driver application, the I ripple current is a Use 2.2µF or higher as a good safe min. LED more important factor when compared to that of the output Rectifier Diode Selection ripple voltage (although the two are directly related). To find the C for a required I ripple use the following A schottky diode is best used here because of the lower OUT LED calculation: forward voltage and the low reverse recovery time. The voltage stress on the diode is the max V and therefore For an output ripple I = 20ma OUT LED(ripple) a diode with a higher rating than max V should be used. OUT ILED(nom) ×Dnom An 80% de-rating is recommended here as well. Eq. (10) C = OUT I ×(R +R )×F Eq. (14) I = I = 0.36A LED(ripple) ADJ LED_total SW DIODE(max) OUT(max) Find the equivalent ac resistance RLED_ac from the Since IIN_AVE(max) occurs when D is at a maximum. datasheet of the LED. This is the inverse slope of the I Eq. (15) P ≈ V × I LED DIODE(max) DIODE DIODE_(max) vs. Vf curve i.e.: A SK35B is used in this example, it’s VDIODE is 0.5V ΔV P ≈ 0.5V × 0.36A = 0.18W Eq. (11) R = f DIODE(max) LED_ac ΔLED MIC3223 Power Losses In this example use R = 0.6Ω for each LED. LED_ac To find the power losses in the MIC3223: If the LEDs are connected in series, multiply R = 0.6Ω LED_ac There is about 6mA input from V into the V pin. by the total number of LEDs. In this example of six LEDs, IN DD we obtain the following: The internal power switch has an RDS of about 170mΩ ON at. R ≡ R = 6 × 0.6Ω = 3.6Ω LED_total dynamic P = V × 6mA + PwrFET Eq. (12) MIC3223 IN Eq. (16) PwrFET = I 2 × R I ×D FET_RMS(max) ds_on_@100° LED(nom) nom C = =1.9μF + V × I × tsw × Fsw OUT I ×(R +R )×F OUT(max) IN_AVE(max) LED(ripple) ADJ LED_total SW R ≈ 160mΩ ds_on_@100° Use 2.2µF or higher. tsw ≈ 30ns is the internal Power FET ON an OFF There is a trade off between the output ripple and the transition time. rising edge of the DIM_IN pulse. This is because between PWM dimming pulses, the converter stops pulsing and ⎛ I 2 ⎞ I = D⎜I 2 + L_PP ⎟ =1.3A COUT will start to discharge. The amount that COUT will SWRMS(max) ⎜ IN_AVE(max) 12 ⎟ discharge depends on the time between PWM Dimming ⎝ ⎠ pluses. At the next DIM_IN pulse, COUT has to be charged PwrFET = 1.3A2 × 160mΩ + 28V × 1.54A × 30ns up to the full output voltage V before the desired LED OUT × 1MHz = 1.6W current flows. P = 8 × 6mA + 1.77W = 1.66W MIC3223 Input Capacitor Snubber The input capacitor is shown in the Typical Application. For superior performance, ceramic capacitors should be A snubber is a damping resistor in series with a DC used because of their low equivalent series resistance blocking capacitor in parallel with the power switch (same (ESR). The input capacitor CIN ripple current is equal to the as across the flyback diode because VOUT is an ac ripple in the inductor. The ripple voltage across the input ground). When the power switch turns off, the drain to capacitor, C is the ESR of C times the inductor ripple. source capacitance and parasitic inductance will cause a IN IN The input capacitor will also bypass the EMI generated by high frequency ringing at the switch node. A snubber the converter as well as any voltage spikes generated by circuit as shown in the application schematic may be the inductance of the input line. For a required V : required if ringing is present at the switch node. A critically IN(ripple) damped circuit at the switch node is where R equals the characteristic impedance of the switch node. Eq. (13) January 2010 16 M9999-011510-A

Micrel, Inc. MIC3223 L parisitic Eq.(17) R = snubber C ds The explanation of the method to find the best R snubber is beyond the scope of this data sheet. Use Rsnubber = 2Ω, ½ watt and Csnubber = 470pf to 1000pf. Figure 10. Simplified Control Loop The power dissipation in the R is: snubber Rsnubber = Csnubber × VOUT2 × FSW Eq. (19) T(s) = Gea(s) × Gvc(s) × H(s) P = 470pF × 28V2 × 1MHz = 0.4W Where snubber R For a LED driver H(s)= ADJ and R +R Power Loss in the L 0.123 W ADJ dynamic Power Loss in the sckottky diode 0.2 W ⎛ ⎛ 1 ⎞⎞ G (s)=g ⎜Z ||⎜R + ⎟⎟ Psnubber 0.4 W ea m⎜ O ⎜ comp sC ⎟⎟ ⎝ ⎝ comp ⎠⎠ MIC3223 Power Loss 1.66 W Eq. (20) Total Losses 2.4W Efficiency 80% V (s) G (s)= OUT Table 2. Major Power Losses VC V (s) CONTROL ⎛ ⎞ Table 2 showing the Power losses in the Design Example. ⎜ sL ⎟( ) 1− 1+sC R OVP - Over Voltage Protection =⎜⎛ 1 ⎟⎞⎜⎛D'ROP⎟⎞⎜⎝ D'2Rdynamic ⎟⎠ OUT ESR ⎝Ri⎠⎝ 2 ⎠ ⎛ sR C ⎞ Set OVP higher than the maximum output voltage by at ⎜1+ dynamic OUT ⎟ ⎜ ⎟ least one Volt. To find the resistor divider values for OVP ⎝ 2 ⎠ use equation 18 and set the OVP = 30V and R = OVP_H Where 100kΩ: V 100kΩ×1.245 R = OUT Is the DC operating point of the converter. Eq. (18) R = = 4.33kΩ OP I OVP_L 30−1.245 LED R is the ac load the converter sees. When the load dymanic Compensation on the converter is a string of LEDs, Rdymanic is the series sum of the R of each LED. LED(ac) R is usually between 0.1Ω to 1Ω per LED. It can be LED_total calculated from the slope of I vs. V plot of the LED. LED f Ri = Ai × Rcs = 0.86Ω Ai = 114 and Rcs ≡ 7.5mΩ; are internal to the ic. The equation for Gvc(s) is theoretical and should give a good idea of where the poles and zeros are located. Figure 9. Current Mode Loop Diagram D'2R D'2R Eq.(20) shows that s= dynamic →f = dynamic Current mode control simplifies the compensation. In L RHPZ 2πL current mode, the complex poles created by the output L is a RHP Zero. The loop bandwidth should be about 1/5 to and C are reduced to a single pole. The explanation for 1/10 of the frequency of R to ensure stability. From HPZ this is beyond the scope of this datasheet, but it’s Equation (20) it is shown that there is only the single pole. generally thought to be because the inductor becomes a 1 1 constant current source and can’t act to change phase. s= →f = and a Zero R C pole 2πR C From the small signal block diagram the loop transfer dynamic OUT dynamic OUT function is: due to the ESR of the output capacitor. 1 1 s= →f = R C ESR 2πR C ESR OUT ESR OUT January 2010 17 M9999-011510-A

Micrel, Inc. MIC3223 This greatly simplifies the compensation. The error amp is a gm type and the gain G (s) is ea One needs only to get a bode plot of the transfer function ⎛ ⎛ 1 ⎞⎞ of the control to output G (s) with a network analyzer Eq. (21) G (s)=g ⎜Z ||⎜R + ⎟⎟ vc ea m⎜ O ⎜ comp sC ⎟⎟ and/or calculate it. From the bode plot find what the gain of ⎝ ⎝ comp ⎠⎠ R G (s) is at f = HPZ . Next design the error amp gain 0.8mA vc 10 gm = V and Zo = 1.2MΩ. G (s) so the loop gain at the cross over frequency T(f ) is ea co 1 f R 0 db where fco= RHPZ or less. The zero is fzero = 2R pC = 1c0o = 1H0P0Z . 10 com comp Error Amp Error Amp Gain and Phase 60 40 Gain E (°) 20 AS 0 H P B) / -20 N (d Phase AI-40 G -60 -80 1.E+02 1.E+03 1.E+04 1.E+05 1.E+06 FREQUENCY (Hz) Set the fco at the mid band where G (f ) = gm × R . At Figure 11. Internal Error Amp and External Compensation ea co comp fzero × 10 the phase boost is near its maximum. Figure 12. Error Amp Transfer Function January 2010 18 M9999-011510-A

Micrel, Inc. MIC3223 Other Applications Figure 13. MIC3223 Typical Application without External PWM Dimming Switch Audio noise 2. Even though the RRC is very short (tens of nanoseconds) the peak currents are high (multiple Audio noise from the output capacitors may exits in a standard boost LED converter. The physical dimensions amperes). These fast current spikes generate EMI (electromagnetic interference). The amount of RRC of ceramic capacitors change with the voltage applied to is related to the die size and internal capacitance of them. During PWM Dimming, the output capacitors in standard converters are subjected to fast voltage and the diode. It is important not to oversize (i.e. not more than the usual de rating) the diode because current transients that may cause the output capacitors the RRC will be needlessly higher. Example: If a 2A to oscillate at the PWM Dimming frequency. This is one reason users may want PWM dimming frequencies diode is needed do not use a higher current rated diode because the RRC will be needlessly higher. If above the audio range. a 25V diode is needed do not use a 100V etc. PCB Layout 3. The high RRC causes a voltage drop on the ground trace of the PCB and if the converter control IC is 1. All typologies of DC-to-DC converters have a referenced to this voltage drop, the output regulation Reverse Recovery Current (RRC) of the flyback will suffer. or (freewheeling) diode. Even a Schottky diode, 4. For good output regulation, it is important to connect which is advertised as having zero RRC, it really the IC’s reference to the same point as the output is not zero. The RRC of the freewheeling diode capacitors to avoid the voltage drop caused by RRC. in a boost converter is even greater than in the This is also called a star connection or single point Buck converter. This is because the output grounding. voltage is higher than the input voltage and the 5. Feedback trace: The high impedance traces of the diode has to charge up to –V during each on- OUT FB should be short. time pulse and then discharge to V during the f off-time. January 2010 19 M9999-011510-A

Micrel, Inc. MIC3223 Evaluation Board Schematic 37V Max 1A LED Driver January 2010 20 M9999-011510-A

Micrel, Inc. MIC3223 Bill of Materials Item Part Number Manufacturer Description Qty GRM319R61E475KA12D muRata(1) C1 C3216X7R1E475M TDK(2) Ceramic Capacitor, 4.7µF, 25V, Size 1206, X7R 1 12063D475KAT2A AVX(3) C2 GRM188R71C273KA01D muRata Ceramic Capacitor, 0.027µF, 6.3V, Size 0603, X7R 1 GRM188R60J106ME47D muRata C3, C7 C1608X5R0J106K TDK Ceramic Capacitor, 10µF, 6.3V, Size 0603, X7R 2 08056D106MAT2A AVX 12105C475KAZ2A AVX C4, C6 Ceramic Capacitor, 4.7µF, 50V, Size 1210, X7R 2 GRM32ER71H475KA88L muRata GRM188R71C473KA01D muRata C5 Ceramic Capacitor, 0.047µF, 6.3V, Size 0603, X7R 1 0603YC473K4T2A AVX C8 GRM188R72A102KA37D muRata Ceramic Capacitor, 1000pF, 100V Size 0603, X7R D1 SK35B MCC(4) Schottky Diode, 3A, 50V (SMB) 1 L1 MSD1260-223ML-LD Coilcraft(6) Inductor, 22µH, 5A 1 R1, R3 CRCW0603100KFKEA Vishay Dale(4) Resistor, 100k, 1%, Size 0603 2 R2 CRCW0603549RFKEA Vishay Dale Resistor, 549Ω, 1%, Size 0603 1 R4 CRCW06033K24FKEA Vishay Dale Resistor, 3.24k, 1%, Size 0603 1 Resistor, 0.56Ω, 1%, 1/2W, Size 1206 R5 CRCW1206R560FKEA Vishay Dale 1 (for .35A LED current Change for different ILED) Stackpole Electronics, R6 RMC 1/4 2 1% R Resistor, 2Ω, 1%, 1/2W, Size 1210 1 Inc.(7) Si2318DS Vishay Siliconix(4) Q1 N-Channel 40V MOSFET 1 AM2340N Analog Power(8) U1 MIC3223 Micrel, Inc.(9) High Power Boost LED Driver with Integrated FET 1 Notes: 1. Murata: www.murata.com. 2. TDK: www.tdk.com. 3. AVX: www.avx.com. 4. Vishay: www.vishay.com. 5. Internacional Rectifier: www.ift.com. 6. Coilcraft: www.coilcraft.com 7. Stackpole Electronics, Inc.: www. 8. Analog Power: www.analogpowerinc.com 8. Micrel, Inc.: www.micrel.com. January 2010 21 M9999-011510-A

Micrel, Inc. MIC3223 PCB Layout Recommendations Top Layer Bottom Layer January 2010 22 M9999-011510-A

Micrel, Inc. MIC3223 Package Information 16-Pin ePad TSSOP (TSE) January 2010 23 M9999-011510-A

Micrel, Inc. MIC3223 Recommended Land Pattern MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems isa Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2009 Micrel, Incorporated. January 2010 24 M9999-011510-A

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