LTC3129-1 15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3µA Quiescent Current FeaTures DescripTion n Regulates V Above, Below or Equal to V The LTC®3129-1 is a high efficiency, 200mA buck-boost OUT IN n Wide V Range: 2.42V to 15V, 1.92V to 15V After DC/DC converter with a wide V and V range. It IN IN OUT Start-Up (Bootstrapped) includes an accurate RUN pin threshold to allow predict- n Fixed Output Voltage with Eight User-Selectable able regulator turn-on and a maximum power point control Settings from 2.5V to 15V (MPPC) capability that ensures maximum power extraction n 200mA Output Current in Buck Mode from non-ideal power sources such as photovoltaic panels. n Single Inductor The LTC3129-1 employs an ultralow noise, 1.2MHz PWM n 1.3µA Quiescent Current switching architecture that minimizes solution footprint by n Programmable Maximum Power Point Control allowing the use of tiny, low profile inductors and ceramic n 1.2MHz Ultralow Noise PWM capacitors. Built-in loop compensation and soft-start n Current Mode Control simplify the design. For high efficiency operation at light n Pin Selectable Burst Mode® Operation loads, automatic Burst Mode operation can be selected, n Up to 95% Efficiency reducing the quiescent current to just 1.3µA. To further n Accurate RUN Pin Threshold reduce part count and improve light load efficiency, the n Power Good Indicator LTC3129-1 includes an internal voltage divider to provide n 10nA Shutdown Current eight selectable fixed output voltages. n Thermally Enhanced 3mm × 3mm QFN and 16-Lead MSOP Packages Additional features include a power good output, less than 10nA of shutdown current and thermal shutdown. applicaTions The LTC3129-1 is available in thermally enhanced n Industrial Wireless Sensor Nodes 3mm × 3mm QFN and 16-lead MSOP packages. For an n Post-Regulator for Harvested Energy adjustable output voltage, see the functionally equivalent n Solar Panel Post-Regulator/Charger LTC3129. n Intrinsically Safe Power Supplies L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks n Wireless Microphones of Linear Technology Corporation. All other trademarks are the property of their respective owners. n Avionics-Grade Wireless Headsets Typical applicaTion Efficiency and Power Loss vs Load 22nF 22nF 10µH 100 1000 90 EFFICIENCY 2.42V TO 1V5IVN VBISNT1 SW1 SW2 BVSOTU2T V5VO UATT 80 100 10µF RUN LTC3129-1 10µF 120000mmAA VVIINN <> VVOOUUTT Y (%) 7600 10 POWER CIENC 50 POWER LOSS LOSS AA OR AAA VCC MPPC PGOOD EFFI 3400 1 (mW) BATTERIES PWM 20 VIN = 2.5V 0.1 VIN = 3.6V VS1 VCC 10 VIN = 5V VS2 VOUT = 5V VIN = 15V 0 0.01 VS3 GND PGND 2.2µF 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) 3129 TA01b 31291 TA01a 31291fc 1 For more information www.linear.com/LTC3129-1

LTC3129-1 absoluTe MaxiMuM raTings (Notes 1, 8) V , V Voltages .....................................–0.3V to 18V V , PWM, MPPC, VS1, VS2, IN OUT CC SW1 DC Voltage ..............................–0.3V to (V + 0.3V) VS3 Voltages ...............................................–0.3V to 6V IN SW2 DC Voltage............................–0.3V to (V + 0.3V) PGOOD Sink Current ..............................................15mA OUT SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V Operating Junction Temperature Range BST1 Voltage .....................(SW1 – 0.3V) to (SW1 + 6V) (Notes 2, 5) ............................................–40°C to 125°C BST2 Voltage .....................(SW2 – 0.3V) to (SW2 + 6V) Storage Temperature Range ..................–65°C to 150°C RUN, PGOOD Voltage .................................–0.3V to 18V MSE Lead Temperature (Soldering, 10 sec) ..........300°C pin conFiguraTion TOP VIEW W1 GND W2 ST2 S P S B TOP VIEW 16 15 14 13 VCC 1 16 VIN BST1 1 12 VOUT RUN 2 15 BST1 MPPC 3 14 SW1 VIN 2 17 11 PGOOD GND 4 17 13 PGND VCC 3 PGND 10 PWM VVSS32 56 PGND 1121 SBWST22 RUN 4 9 VS1 VS1 7 10 VOUT PWM 8 9 PGOOD 5 6 7 8 C D 3 2 MSE PACKAGE PP GN VS VS 16-LEAD PLASTIC MSOP M UD PACKAGE TJMAX = 125°C, θJC = 10°C/W, θJA = 40°C/W (NOTE 6) EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB 16-LEAD (3mm × 3mm) PLASTIC QFN TJMAX = 125°C, θJC = 7.5°C/W, θJA = 68°C/W (NOTE 6) EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB orDer inForMaTion LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3129EUD-1#PBF LTC3129EUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C LTC3129IUD-1#PBF LTC3129IUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C LTC3129EMSE-1#PBF LTC3129EMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C LTC3129IMSE-1#PBF LTC3129IMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 31291fc 2 For more information www.linear.com/LTC3129-1

LTC3129-1 elecTrical characTerisTics The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at T = 25°C (Note 2). Unless otherwise noted, V = 12V, V = 5V. A IN OUT PARAMETER CONDITIONS MIN TYP MAX UNITS V Start-Up Voltage l 2.25 2.42 V IN Input Voltage Range V > 2.42V (Back-Driven) l 1.92 15 V CC V UVLO Threshold (Rising) V > 2.42V (Back-Driven) l 1.8 1.9 2.0 V IN CC V UVLO Hysteresis l 80 100 130 mV IN V Voltages VS1 = VS2 = VS3 = 0V l 2.425 2.5 2.575 V OUT VS1 = V , VS2 = VS3 = 0V l 3.2175 3.3 3.383 V CC VS2 = V , VS1 = VS3 = 0V l 3.998 4.1 4.203 V CC VS1 = VS2 = V , VS3 = 0V l 4.875 5.0 5.125 V CC VS1 = VS2 = 0V, VS3 = V l 6.727 6.9 7.073 V CC VS2 = 0V, VS1 = VS3 = V l 7.995 8.2 8.405 V CC VS1 = 0V, VS2 = VS3 = V l 11.64 12 12.40 V CC VS1 = VS2 = VS3 = V l 14.50 15.0 15.50 V CC Quiescent Current (V ) – Shutdown RUN = 0V, Including Switch Leakage 10 100 nA IN Quiescent Current (V ) UVLO Either V or V Below Their UVLO Threshold, or 1.9 3 µA IN IN CC RUN Below the Threshold to Enable Switching Quiescent Current – Burst Mode Operation Measured on V , V > V 1.3 2.0 µA IN OUT REG PWM = 0V, RUN = V IN N-Channel Switch Leakage on V and V SW1 = 0V, V = 15V 10 50 nA IN OUT IN SW2 = 0V, V = 15V OUT RUN = 0V N-Channel Switch On-Resistance V = 4V 0.75 Ω CC Inductor Average Current Limit V > UV Threshold (Note 4) l 220 275 350 mA OUT V < UV Threshold (Note 4) l 80 130 200 mA OUT Inductor Peak Current Limit (Note 4) l 400 500 680 mA Maximum Boost Duty Cycle V < V as Set by VS1-VS3. Percentage of l 85 89 95 % OUT REG Period SW2 is Low in Boost Mode (Note 7) Minimum Duty Cycle V > V as Set by VS1-VS3. Percentage of l 0 % OUT REG Period SW1 is High in Buck Mode (Note 7) Switching Frequency PWM = V l 1.0 1.2 1.4 MHz CC SW1 and SW2 Minimum Low Time (Note 3) 90 ns MPPC Voltage l 1.12 1.175 1.22 V MPPC Input Current MPPC = 5V 1 10 nA RUN Threshold to Enable V l 0.5 0.9 1.15 V CC RUN Threshold to Enable Switching (Rising) V > 2.4V l 1.16 1.22 1.28 V CC RUN (Switching) Threshold Hysteresis 50 80 120 mV RUN Input Current RUN = 15V 1 10 nA VS1, VS2, VS3 Input High l 1.2 V VS1, VS2, VS3 Input Low l 0.4 V VS1, VS2, VS3 Input Current VS1, VS2, VS3 = V = 5V 1 10 nA CC PWM Input High l 1.6 V PWM Input Low l 0.5 V PWM Input Current PWM = 5V 0.1 1 µA Soft-Start Time 3 ms V Voltage V > 4.85V l 3.4 4.1 4.7 V CC IN V Dropout Voltage (V – V ) V = 3.0V, Switching 35 60 mV CC IN CC IN V = 2.0V (V in UVLO) 0 2 mV IN CC 31291fc 3 For more information www.linear.com/LTC3129-1

LTC3129-1 elecTrical characTerisTics The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at T = 25°C (Note 2). Unless otherwise noted, V = 12V, V = 5V. A IN OUT PARAMETER CONDITIONS MIN TYP MAX UNITS V UVLO Threshold (Rising) l 2.1 2.25 2.42 V CC V UVLO Hysteresis 60 mV CC V Current Limit V = 0V l 4 20 60 mA CC CC V Back-Drive Voltage (Maximum) l 5.5 V CC V Input Current (Back-Driven) V = 5.5V (Switching) 2 4 mA CC CC V Leakage to V if V >V V = 5.5V, V = 1.8V, Measured on V –27 µA CC IN CC IN CC IN IN V UV Threshold (Rising) l 0.95 1.15 1.35 V OUT V UV Hysteresis 150 mV OUT V Current – Shutdown RUN = 0V, V = 15V Including Switch Leakage 10 100 nA OUT OUT V Current – Sleep PWM = 0V, V ≥ V V /27 µA OUT OUT REG OUT V Current – Active PWM = V , V = 15V (Note 4) 5 9 µA OUT CC OUT PGOOD Threshold, Falling Referenced to Programmed V Voltage –5.5 –7.5 –10 % OUT PGOOD Hysteresis Referenced to Programmed V Voltage 2.5 % OUT PGOOD Voltage Low I = 1mA 250 300 mV SINK PGOOD Leakage PGOOD = 15V 1 50 nA Note 1: Stresses beyond those listed under Absolute Maximum Ratings Note 3: Specification is guaranteed by design and not 100% tested in may cause permanent damage to the device. Exposure to any Absolute production. Maximum Rating condition for extended periods may affect device Note 4: Current measurements are made when the output is not switching. reliability and lifetime. Note 5: This IC includes overtemperature protection that is intended Note 2: The LTC3129-1 is tested under pulsed load conditions such to protect the device during momentary overload conditions. Junction that TJ ≈ TA. The LTC3129E-1 is guaranteed to meet specifications temperature will exceed 125°C when overtemperature protection is active. from 0°C to 85°C junction temperature. Specifications over the –40°C Continuous operation above the specified maximum operating junction to 125°C operating junction temperature range are assured by design, temperature may result in device degradation or failure. characterization and correlation with statistical process controls. The Note 6: Failure to solder the exposed backside of the package to the PC LTC3129I-1 is guaranteed over the full –40°C to 125°C operating junction board ground plane will result in a much higher thermal resistance. temperature range. The junction temperature (T , in °C) is calculated from J Note 7: Switch timing measurements are made in an open-loop test the ambient temperature (T , in °C) and power dissipation (P , in watts) A D configuration. Timing in the application may vary somewhat from these according to the formula: values due to differences in the switch pin voltage during non-overlap TJ = TA + (PD • θJA), durations when switch pin voltage is influenced by the magnitude and where θJA (in °C/W) is the package thermal impedance. duration of the inductor current. Note that the maximum ambient temperature consistent with these Note 8: Voltage transients on the switch pin(s) beyond the DC limits specifications is determined by specific operating conditions in specified in the Absolute Maximum Ratings are non-disruptive to normal conjunction with board layout, the rated thermal package thermal operation when using good layout practices as described elsewhere in the resistance and other environmental factors. data sheet and Application Notes and as seen on the product demo board. 31291fc 4 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A Efficiency, V = 2.5V Power Loss, V = 2.5V Efficiency, V = 3.3V OUT OUT OUT 100 1000 100 90 BURST 90 BURST 80 100 PWM 80 %) 70 mW) %) 70 NCY ( 5600 OSS ( 10 NCY ( 5600 EFFICIE 4300 PWVMIN = 2.5V POWER L 1 VIN = 2.5V EFFICIE 4300 PWVMIN = 2.5V 20 VVIINN == 35.V6V 0.1 BURST VVIINN == 35.V6V 20 VVIINN == 35.V6V 10 VIN = 10V VIN = 10V 10 VIN = 10V VIN = 15V VIN = 15V VIN = 15V 0 0.01 0 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) 31291 G01 31291 G02 31291 G03 Power Loss, VOUT = 3.3V Efficiency, VOUT = 4.1V Power Loss, VOUT = 4.1V 1000 100 1000 BURST 90 100 PWM 80 100 PWM W) 70 W) m %) m S ( 10 Y ( 60 S ( 10 S C S O N 50 O R L CIE PWM R L WE 1 FFI 40 WE 1 BURST PO VIN = 2.5V E 30 VIN = 2.5V PO VIN = 2.5V 0.1 BURST VIN = 3.6V 20 VIN = 3.6V 0.1 VIN = 3.6V VIN = 5V VIN = 5V VIN = 5V VIN = 10V 10 VIN = 10V VIN = 10V VIN = 15V VIN = 15V VIN = 15V 0.01 0 0.01 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) 31291 G04 31291 G04a 31291 G04b Efficiency, V = 5V Power Loss, V = 5V Efficiency, V = 6.9V OUT OUT OUT 100 1000 100 BURST 90 BURST 90 80 100 PWM 80 70 W) 70 %) m %) Y ( 60 S ( 10 Y ( 60 C S C N 50 O N 50 CIE PWM R L CIE PWM FFI 40 WE 1 FFI 40 E 30 VIN = 2.5V PO VIN = 2.5V E 30 VIN = 2.5V 20 VIN = 3.6V 0.1 BURST VIN = 3.6V 20 VIN = 3.6V VIN = 5V VIN = 5V VIN = 5V 10 VIN = 10V VIN = 10V 10 VIN = 10V VIN = 15V VIN = 15V VIN = 15V 0 0.01 0 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) 31291 G05 31291 G06 31291 G06a 31291fc 5 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A Power Loss, V = 6.9V Efficiency, V = 8.2V Power Loss, V = 8.2V OUT OUT OUT 1000 100 1000 BURST 90 100 PWM 80 100 PWM W) 70 W) m %) m S ( 10 Y ( 60 S ( 10 S C S O N 50 O R L CIE PWM R L WE 1 BURST FFI 40 WE 1 BURST PO VIN = 2.5V E 30 VIN = 2.5V PO VIN = 2.5V 0.1 VIN = 3.6V 20 VIN = 3.6V 0.1 VIN = 3.6V VIN = 5V VIN = 5V VIN = 5V VIN = 10V 10 VIN = 10V VIN = 10V VIN = 15V VIN = 15V VIN = 15V 0.01 0 0.01 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) 31291 G06b 31291 G06c 31291 G06d Efficiency, V = 12V Power Loss, V = 12V Efficiency, V = 15V OUT OUT OUT 100 1000 100 BURST BURST 90 90 PWM 80 100 80 70 W) 70 %) m %) Y ( 60 S ( 10 Y ( 60 C S C N 50 O N 50 FFICIE 40 PWM WER L 1 BURST FFICIE 40 PWM E O E 30 VIN = 2.5V P VIN = 2.5V 30 VIN = 2.5V 20 VIN = 3.6V 0.1 VIN = 3.6V 20 VIN = 3.6V VIN = 5V VIN = 5V VIN = 5V 10 VIN = 10V VIN = 10V 10 VIN = 10V VIN = 15V VIN = 15V VIN = 15V 0 0.01 0 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) 31291 G07 31291 G08 31291 G09 Maximum Output Current No Load Input Current Power Loss, V = 15V vs V and V vs V and V (PWM = 0V) OUT IN OUT IN OUT 1000 250 5 VOUT = 2.5V VOUT = 3.3V 100 PWM 200 4 VVOOUUTT == 45.V1V VOUT = 6.9V W) VOUT = 8.2V ER LOSS (m 101 BURST I (mA)OUT115000 VVOOUUTT == 23..53VV I (µA)IN 32 VVOOUUTT == 1125VV W VOUT = 4.1V PO VIN = 2.5V VOUT = 5V 0.1 VVIINN == 35.V6V 50 VVOOUUTT == 68..92VV 1 VIN = 10V VOUT = 12V VIN = 15V VOUT = 15V 0.01 0 0 0.01 0.1 1 10 100 1000 2 3 4 5 6 7 8 9 10 11 12 13 14 15 2 4 6 8 10 12 14 16 OUTPUT CURRENT (mA) VIN (V) VIN (V) 31291 G10 31291 G11 31291 G12 31291fc 6 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A Burst Mode Threshold Output Voltage vs Temperature vs V and V Switch R vs Temperature (Normalized to 25°C) IN OUT DS(ON) 80 1.3 1.0 VCC = 2.5V 70 1.2 VCC = 3V VCC = 4V 60 1.1 VCC = 5V 0.5 %) LOAD (mA) 435000 VVVOOOUUUTTT === 234...531VVV R (Ω)DS(ON)1000....0879 CHANGE IN V (OUT 0 20 VOUT = 5V 0.6 –0.5 VOUT = 6.9V 10 VOUT = 8.2V 0.5 VOUT = 12V 0 VOUT = 15V 0.4 –1.0 2 4 6 8 10 12 14 16 –45 –20 5 30 55 80 105 130 –45 –20 5 30 55 80 105 130 VIN (V) TEMPERATURE (°C) TEMPERATURE (°C) 3129 G13 31291 G14 31291 G15 Accurate RUN Threshold Average Input Current Limit Maximum Output vs Temperature vs Temperature (Normalized to 25°C) vs MPPC Voltage (Normalized to 25°C) 2 %)100 %) 15 HRESHOLD (%) 1 NPUT CURRENT ( 97680000 UTPUT CURRENT ( 105 N T 0 LL I 50 M O 0 CHANGE IN RU–1 CENTAGE OF FU 432000 GE IN MAXIMU––150 R 10 N E A P H –2 0 C–15 –45 –20 5 30 55 80 105 130 1.131.135 1.14 1.1451.15 1.155 1.161.165 1.17 –45 –20 5 30 55 80 105 130 TEMPERATURE (°C) MPPC PIN VOLTAGE (V) TEMPERATURE (°C) 31291 G17 31291 G18 31291 G19 VCC Dropout Voltage vs Temperature VCC Dropout Voltage vs VIN Fixed Frequency PWM (PWM Mode, Switching) (PWM Mode, Switching) Waveforms 60 60 SW2 50 50 5V/DIV mV) 40 mV) 40 5VS/DWIV1 T ( T ( U 30 U 30 O O P P RO RO IL D 20 D 20 200mA/DIV 10 10 500ns/DIV 31291 G22 L = 10µH 0 0 VIN = 7V –45 –20 5 30 55 80 105 130 2 2.25 2.5 2.75 3 3.25 3.5 3.75 4 VOUT = 5V TEMPERATURE (°C) VIN (V) IOUT = 200mA 31291 G20 31291 G21 31291fc 7 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A Fixed Frequency Ripple on V Burst Mode Waveforms Burst Mode Ripple on V OUT OUT SW1 VOUT 5V/DIV 100mVV/ODUIVT 20mV/DIV SW2 5V/DIV IL 200mA/DIV IL IL 200mA/DIV 100mA/DIV 200ns/DIV 31291 G23 50µs/DIV 31291 G24 100µs/DIV 31291 G25 L = 10µH L = 10µH L = 10µH VIN = 7V VIN = 7V VIN = 7V VOUT = 5V VOUT = 5V VOUT = 5V IOUT = 200mA IOUT = 5mA IOUT = 5mA COUT = 10µF COUT = 22µF COUT = 22µF Step Load Transient Response in Step Load Transient Response in Start-Up Waveforms Fixed Frequency Burst Mode Operation VOUT 5V/DIV VOUT VOUT VCC 100mV/DIV 100mV/DIV 5V/DIV RUN 5V/DIV IVOUT IVOUT IVIN 100mA/DIV 100mA/DIV 200mA/DIV 1ms/DIV 31291 G26 500µs/DIV 31291 G27 500µs/DIV 31291 G28 VIN = 7V L = 10µH L = 10µH VOUT = 5V VIN = 7V VIN = 7V IOUT = 50mA VOUT = 5V VOUT = 5V COUT = 22µA COUT = 10µF COUT = 22µF IOUT = 50mA to 150mA STEP IOUT = 5mA to 125mA STEP PGOOD Response to a Drop On V MPPC Response to a Step Load OUT VOUT PGOOD 2V/DIV 2V/DIV VIN 2V/DIV VOUT 2V/DIV IVOUT 100mA/DIV 1ms/DIV 31291 G29 2ms/DIV 31291 G30 VOUT = 5V VIN = 5VOC VMPPC SET TO 3.5V CIN = 22µF, RIN = 10Ω, VOUT = 5V, COUT = 22µF IOUT = 25mA to 125mA STEP 31291fc 8 For more information www.linear.com/LTC3129-1

LTC3129-1 pin FuncTions (QFN/MSOP) BST1 (Pin 1/Pin 15): Boot-Strapped Floating Supply for VS3 (Pin 7/Pin 5): Output Voltage Select Pin. Connect this High Side NMOS Gate Drive. Connect to SW1 through a pin to ground or V to program the output voltage (see CC 22nF capacitor, as close to the part as possible. The value Table 1). This pin should not float or go below ground. is not critical. Any value from 4.7nF to 47nF may be used. If this pin is externally driven above V , a 1M resistor CC should be added in series. V (Pin 2/Pin 16): Input Voltage for the Converter. Connect IN a minimum of 4.7µF ceramic decoupling capacitor from VS2 (Pin 8/Pin 6): Output Voltage Select Pin. Connect this this pin to the ground plane, as close to the pin as possible. pin to ground or V to program the output voltage (see CC Table 1). This pin should not float or go below ground. V (Pin 3/Pin 1): Output Voltage of the Internal Voltage CC Regulator. This is the supply pin for the internal circuitry. VS1 (Pin 9/Pin 7): Output Voltage Select Pin. Connect this Bypass this output with a minimum of 2.2µF ceramic ca- pin to ground or V to program the output voltage (see CC pacitor close to the pin. This pin may be back-driven by Table 1). This pin should not float or go below ground. an external supply, up to a maximum of 5.5V. PWM (Pin 10/Pin 8): Mode Select Pin. RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull PWM = Low (ground): Enables automatic Burst Mode this pin above 1.1V to enable the V regulator and above CC operation. 1.28V to enable the converter. Connecting this pin to a resistor divider from VIN to ground allows programming a PWM = High (tie to VCC): Fixed frequency PWM V start threshold higher than the 1.8V (typical) V UVLO operation. IN IN threshold. In this case, the typical V turn-on threshold is IN This pin should not be allowed to float. It has internal 5M determined by V = 1.22V • [1+(R3/Pin R4)] (see Figure 2). IN pull-down resistor. MPPC (Pin 5/Pin 3): Maximum Power Point Control PGOOD (Pin 11/Pin 9): Open drain output that pulls to Programming Pin. Connect this pin to a resistor divider ground when FB drops too far below its regulated voltage. from V to ground to enable the MPPC functionality. IN Connect a pull-up resistor from this pin to a positive sup- If the V load is greater than what the power source OUT ply. This pin can sink up to the absolute maximum rating can provide, the MPPC will reduce the inductor current of 15mA when low. Refer to the Operation section of the to regulate V to a voltage determined by: V = 1.175V IN IN data sheet for more detail. • [1 + (R5/R6)] (see Figure 3). By setting the V regula- IN V (Pin 12/Pin 10): Output voltage of the converter, set tion voltage appropriately, maximum power transfer from OUT by the VS1-VS3 programming pins according to Table 1. the limited source is assured. Note this pin is very noise Connect a minimum value of 4.7µF ceramic capacitor from sensitive, therefore minimize trace length and stray capaci- this pin to the ground plane, as close to the pin as possible. tance. Please refer to the Applications Information section for more detail on programming the MPPC for different BST2 (Pin 13/Pin 11): Boot-Strapped Floating Supply for sources. If this function is not needed, tie the pin to VCC. High Side NMOS Gate Drive. Connect to SW2 through a 22nF capacitor, as close to the part as possible. The value GND (Pin 6/Pin 4): Signal Ground. Provide a short direct is not critical. Any value from 4.7nF to 47nF may be used PCB path between GND and the ground plane where the exposed pad is soldered. SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of the inductor. Keep PCB trace lengths as short and wide as possible to reduce EMI. 31291fc 9 For more information www.linear.com/LTC3129-1

LTC3129-1 pin FuncTions (QFN/MSOP) PGND (Pin 15/Pin 13, Exposed Pad Pin 17/Pin 17): Power Table 1. V Program Settings OUT Ground. Provide a short direct PCB path between PGND VS3 PIN VS2 PIN VS1 PIN V OUT and the ground plane. The exposed pad must also be 0 0 0 2.5V soldered to the PCB ground plane. It serves as a power 0 0 V 3.3V CC ground connection, and as a means of conducting heat 0 V 0 4.1V CC away from the die. 0 V V 5V CC CC SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of V 0 0 6.9V CC the inductor. Keep PCB trace lengths as short and wide V 0 V 8.2V CC CC as possible to reduce EMI. V V 0 12V CC CC V V V 15V CC CC CC block DiagraM BST1 SW1 SW2 BST2 VIN VIN VCC VREF START LDO VCC_GD VOUT DRIVER A VOUT VCC 4.1V VCC VCC ISENSE D DRIVER START VREF 1V.R1E7F5V DRIVER B ISENSE VREF_GD C DRIVER VS1 RUN 0.9V +– START DRV_B DRV_C VS2 VSINOEPLUUETCTST DRV_A DRV_D VS3 + SD 1.22V – ISENSE + UV – VIN 500mA – ILIM + 1.1V 1.175V LOGIC – ENABLE ISENSE UVLO + ISENSE – IZERO – VC – FB 20mA + PWM + + + 1.175V THERMAL – SHUTDOWN RESET SOFT-START MPPC + OSC 1.175V – – PGOOD PWM 600mV + – 5M – SSLLEEEEPP CLAMP –7.5% + 100mV + GND PGND 31291 BD 31291fc 10 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion INTRODUCTION voltage range, 1.3µA Burst Mode current and program- mable RUN and MPPC pins, the LTC3129-1 is well suited The LTC3129-1 is a 1.3µA quiescent current, monolithic, for many diverse applications. current mode, buck-boost DC/DC converter that can operate over a wide input voltage range of 1.92V to 15V and provide up to 200mA to the load. Eight fixed, user-programmable PWM MODE OPERATION output voltages can be selected using the three digital If the PWM pin is high or if the load current on the con- programming pins. Internal, low R N-channel power DS(ON) verter is high enough to command PWM mode operation switches reduce solution complexity and maximize effi- with PWM low, the LTC3129-1 operates in a fixed 1.2MHz ciency. A proprietary switch control algorithm allows the PWM mode using an internally compensated average buck-boost converter to maintain output voltage regulation current mode control loop. PWM mode minimizes output with input voltages that are above, below or equal to the voltage ripple and yields a low noise switching frequency output voltage. Transitions between the step-up or step- spectrum. A proprietary switching algorithm provides down operating modes are seamless and free of transients seamless transitions between operating modes and and sub-harmonic switching, making this product ideal eliminates discontinuities in the average inductor cur- for noise sensitive applications. The LTC3129-1 operates rent, inductor ripple current and loop transfer function at a fixed nominal switching frequency of 1.2MHz, which throughout all modes of operation. These advantages provides an ideal trade-off between small solution size and result in increased efficiency, improved loop stability and high efficiency. Current mode control provides inherent lower output voltage ripple in comparison to the traditional input line voltage rejection, simplified compensation and buck-boost converter. rapid response to load transients. Figure 1 shows the topology of the LTC3129-1 power stage Burst Mode capability is also included in the LTC3129-1 which is comprised of four N-channel DMOS switches and and is user-selected via the PWM input pin. In Burst Mode their associated gate drivers. In PWM mode operation operation, the LTC3129-1 provides exceptional efficiency at both switch pins transition on every cycle independent of light output loading conditions by operating the converter the input and output voltages. In response to the internal only when necessary to maintain voltage regulation. The control loop command, an internal pulse width modulator Burst Mode quiescent current is a miserly 1.3µA. At higher generates the appropriate switch duty cycle to maintain loads, the LTC3129-1 automatically switches to fixed fre- regulation of the output voltage. quency PWM mode when Burst Mode operation is selected. (Please refer to the Typical Performance Characteristic CBST1 CBST2 L curves for the mode transition point at different input and output voltages). If the application requires extremely low BST1 VIN SW1 SW2 VOUT BST2 noise, continuous PWM operation can also be selected VCC VCC via the PWM pin. A D A MPPC (maximum power point control) function is also provided that allows the input voltage to the converter to VCC VCC be servo’d to a programmable point for maximum power B C when operating from various non-ideal power sources such as photovoltaic cells. The LTC3129-1 also features PGND PGND LTC3129-1 an accurate RUN comparator threshold with hysteresis, 31291 F01 allowing the buck-boost DC/DC converter to turn on and Figure 1. Power Stage Schematic off at user-selected V voltage thresholds. With a wide IN 31291fc 11 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion When stepping down from a high input voltage to a lower ramps, and the comparator outputs are used to control output voltage, the converter operates in buck mode and the duty cycle of the switch pins on a cycle-by-cycle basis. switch D remains on for the entire switching cycle except The voltage error amplifier monitors the output voltage, for the minimum switch low duration (typically 90ns). Dur- V through the internal voltage divider and makes adjust- OUT ing the switch low duration, switch C is turned on which ments to the current command as necessary to maintain forces SW2 low and charges the flying capacitor, C . BST2 regulation. The voltage error amplifier therefore controls This ensures that the switch D gate driver power supply the outer voltage regulation loop. The average current rail on BST2 is maintained. The duty cycle of switches A amplifier makes adjustments to the inductor current as and B are adjusted to maintain output voltage regulation directed by the voltage error amplifier output via V and is C in buck mode. commonly referred to as the inner current loop amplifier. If the input voltage is lower than the output voltage, the The average current mode control technique is similar to converter operates in boost mode. Switch A remains on peak current mode control except that the average current for the entire switching cycle except for the minimum amplifier, by virtue of its configuration as an integrator, switch low duration (typically 90ns). During the switch controls average current instead of the peak current. This low duration, switch B is turned on which forces SW1 difference eliminates the peak to average current error low and charges the flying capacitor, C . This ensures BST1 inherent to peak current mode control, while maintaining that the switch A gate driver power supply rail on BST1 is most of the advantages inherent to peak current mode maintained. The duty cycle of switches C and D are adjusted control. to maintain output voltage regulation in boost mode. Average current mode control requires appropriate com- Oscillator pensation for the inner current loop, unlike peak current mode control. The compensation network must have high The LTC3129-1 operates from an internal oscillator with a DC gain to minimize errors between the actual and com- nominal fixed frequency of 1.2MHz. This allows the DC/DC manded average current level, high bandwidth to quickly converter efficiency to be maximized while still using small change the commanded current level following transient external components. load steps and a controlled mid-band gain to provide a form of slope compensation unique to average current Current Mode Control mode control. The compensation components required The LTC3129-1 utilizes average current mode control for to ensure proper operation have been carefully selected the pulse width modulator. Current mode control, both and are integrated within the LTC3129-1. average and the better known peak method, enjoy some benefits compared to other control methods including: Inductor Current Sense and Maximum Output Current simplified loop compensation, rapid response to load As part of the current control loop required for current transients and inherent line voltage rejection. mode control, the LTC3129-1 includes a pair of current Referring to the Block Diagram, a high gain, internally sensing circuits that measure the buck-boost converter compensated transconductance amplifier monitors VOUT inductor current. through an internal voltage divider. The error amplifier out- The voltage error amplifier output, V , is internally clamped put is used by the current mode control loop to command C to a nominal level of 0.6V. Since the average inductor the appropriate inductor current level. The inverting input current is proportional to V , the 0.6V clamp level sets of the internally compensated average current amplifier is C the maximum average inductor current that can be pro- connected to the inductor current sense circuit. The aver- grammed by the inner current loop. Taking into account age current amplifier’s output is compared to the oscillator the current sense amplifier’s gain, the maximum average 31291fc 12 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion inductor current is approximately 275mA (typical). In Overload Current Limit and I Comparator ZERO buck mode, the output current is approximately equal to The internal current sense waveform is also used by the the inductor current I . L peak overload current (I ) and zero current (I ) com- PEAK ZERO I ≈ I • 0.89 parators. The I current comparator monitors I OUT(BUCK) L PEAK SENSE and turns off switch A if the inductor current level exceeds The 90ns SW1/SW2 forced low time on each switching its maximum internal threshold, which is approximately cycle briefly disconnects the inductor from V and V OUT IN 500mA. An inductor current level of this magnitude will resulting in about 11% less output current in either buck occur during a fault, such as an output short-circuit, or or Boost mode for a given inductor current. In boost mode, during large load or input voltage transients. the output current is related to average inductor current and duty cycle by: The LTC3129-1 features near discontinuous inductor current operation at light output loads by virtue of the I ≈ I • (1 – D) • Efficiency OUT(BOOST) L I comparator circuit. By limiting the reverse current ZERO where D is the converter duty cycle. magnitude in PWM mode, a balance between low noise operation and improved efficiency at light loads is achieved. Since the output current in boost mode is reduced by the The I comparator threshold is set near the zero current duty cycle (D), the output current rating in buck mode is ZERO level in PWM mode, and as a result, the reverse current always greater than in boost mode. Also, because boost magnitude will be a function of inductance value and out- mode operation requires a higher inductor current for a put voltage due to the comparator's propagation delay. In given output current compared to buck mode, the efficiency in boost mode will be lower due to higher I 2 • R general, higher output voltages and lower inductor values L DS(ON) will result in increased reverse current magnitude. losses in the power switches. This will further reduce the output current capability in boost mode. In either operating In automatic Burst Mode operation (PWM pin low), the mode, however, the inductor peak-to-peak ripple current I comparator threshold is increased so that reverse ZERO does not play a major role in determining the output cur- inductor current does not normally occur. This maximizes rent capability, unlike peak current mode control. efficiency at very light loads. With peak current mode control, the maximum output current capability is reduced by the magnitude of inductor Burst Mode OPERATION ripple current because the peak inductor current level is the When the PWM pin is held low, the LTC3129-1 is con- control variable, but the average inductor current is what figured for automatic Burst Mode operation. As a result, determines the output current. The LTC3129-1 measures the buck-boost DC/DC converter will operate with normal and controls average inductor current, and therefore, the continuous PWM switching above a predetermined mini- inductor ripple current magnitude has little effect on the mum output load and will automatically transition to power maximum current capability in contrast to an equivalent saving Burst Mode operation below this output load level. peak current mode converter. Under most conditions in Note that if the PWM pin is low, reverse inductor current is buck mode, the LTC3129-1 is capable of providing a mini- not allowed at any load. Refer to the Typical Performance mum of 200mA to the load. In boost mode, as described Characteristics section of this data sheet to determine the previously, the output current capability is related to the Burst Mode transition threshold for various combinations boost ratio or duty cycle (D). For example, for a 3.6V V IN of V and V . If PWM is low, at light output loads, the IN OUT to 5V output application, the LTC3129-1 can provide up to 150mA to the load. Refer to the Typical Performance Characteristics section for more detail on output current capability. 31291fc 13 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion LTC3129-1 will go into a standby or sleep state when the protection to safeguard against accidental short-circuiting output voltage achieves its nominal regulation level. The of the V rail. CC sleep state halts PWM switching and powers down all nonessential functions of the IC, significantly reducing the Undervoltage Lockout (UVLO) quiescent current of the LTC3129-1 to just 1.3µA typical. There are two undervoltage lockout (UVLO) circuits within This greatly improves overall power conversion efficiency the LTC3129-1 that inhibit switching; one that monitors V IN when the output load is light. Since the converter is not and another that monitors V . Either UVLO will disable CC operating in sleep, the output voltage will slowly decay operation of the internal power switches and keep other at a rate determined by the output load resistance and IC functions in a reset state if either V or V are below IN CC the output capacitor value. When the output voltage has their respective UVLO thresholds. decayed by a small amount, the LTC3129-1 will wake and The V UVLO comparator has a falling voltage threshold resume normal PWM switching operation until the volt- IN of 1.8V (typical). If V falls below this level, IC operation age on V is restored to the previous level. If the load IN OUT is disabled until V rises above 1.9V (typical), as long as is very light, the LTC3129-1 may only need to switch for IN the V voltage is above its UVLO threshold. a few cycles to restore V and may sleep for extended CC OUT periods of time, significantly improving efficiency. If the The V UVLO has a falling voltage threshold of 2.19V CC load is suddenly increased above the burst transition (typical). If the V voltage falls below this threshold, IC CC threshold, the part will automatically resume continuous operation is disabled until V rises above 2.25V (typical) CC PWM operation until the load is once again reduced. as long as V is above its nominal UVLO threshold level. IN Note that Burst Mode operation is inhibited until soft-start Depending on the particular application, either of these is done, the MPPC pin is greater than 1.175V and VOUT UVLO thresholds could be the limiting factor affecting the has reached regulation. minimum input voltage required for operation. Because the V regulator uses V for its power input, the minimum CC IN Soft-Start input voltage required for operation is determined by the The LTC3129-1 soft-start circuit minimizes input current VCC minimum voltage, as input voltage (VIN) will always transients and output voltage overshoot on initial power up. be higher than VCC in the normal (non-bootstrapped) The required timing components for soft-start are internal configuration. Therefore, the minimum VIN for the part to the LTC3129-1 and produce a nominal soft-start dura- to start up is 2.25V (typical). tion of approximately 3ms. The internal soft-start circuit In applications where V is bootstrapped (powered CC slowly ramps the error amplifier output, V . In doing so, C through a Schottky diode by either V or an auxiliary OUT the current command of the IC is also slowly increased, power rail), the minimum input voltage for operation will starting from zero. It is unaffected by output loading or be limited only by the V UVLO threshold (1.8V typical). IN output capacitor value. Soft-start is reset by the UVLO on Please note that if the bootstrap voltage is derived from both V and V , the RUN pin and thermal shutdown. IN CC the LTC3129-1 V and not an independent power rail, OUT then the minimum input voltage required for initial start-up V Regulator CC is still 2.25V (typical). An internal low dropout regulator (LDO) generates a nomi- Note that if either V or V are below their UVLO IN CC nal 4.1V V rail from V . The V rail powers the internal CC IN CC thresholds, or if RUN is below its accurate threshold of control circuitry and the gate drivers of the LTC3129-1. The 1.22V (typical), then the LTC3129-1 will remain in a soft V regulator is disabled in shutdown to reduce quiescent CC shutdown state, where the V quiescent current will be IN current and is enabled by raising the RUN pin above its only 1.9µA typical. logic threshold. The V regulator includes current-limit CC 31291fc 14 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion V Undervoltage OUT LTC3129-1 There is also an undervoltage comparator that monitors VIN ACCURATE THRESHOLD the output voltage. Until V reaches 1.15V (typical), the 1.22V – OUT ENABLE SWITCHING R3 + average current limit is reduced by a factor of two. This RUN reduces power dissipation in the device in the event of a shorted output. In addition, N-channel switch D, which R4 + ENABLE LDO AND feeds VOUT, will be disabled until VOUT exceeds 1.15V. 0.9V – CONTROL CIRCUITS LOGIC THRESHOLD RUN Pin Comparator 31291 F02 In addition to serving as a logic level input to enable cer- Figure 2. Accurate RUN Pin Comparator tain functions of the IC, the RUN pin includes an accurate internal comparator that allows it to be used to set custom Note that once RUN is above 0.9V typical, the quiescent rising and falling ON/OFF thresholds with the addition of input current on VIN (or VCC if back-driven) will increase to an optional external resistor divider. When RUN is driven about 1.9µA typical until the VIN and VCC UVLO thresholds above its logic threshold (0.9V typical), the V regulator are satisfied. CC is enabled, which provides power to the internal control The converter is enabled when the voltage on RUN exceeds circuitry of the IC. If the voltage on RUN is increased 1.22V (nominal). Therefore, the turn on voltage threshold further so that it exceeds the RUN comparator’s accurate on V is given by: IN analog threshold (1.22V typical), all functions of the buck- V = 1.22V • (1 + R3/R4) boost converter will be enabled and a start-up sequence IN(TURN-ON) will ensue (assuming the VIN and VCC UVLO thresholds The RUN comparator includes a built-in hysteresis of are satisfied). approximately 80mV, so that the turn off threshold will be 1.14V. If RUN is brought below the accurate comparator threshold, the buck-boost converter will inhibit switching, but the VCC There may be cases due to PCB layout, very large value regulator and control circuitry will remain powered unless resistors for R3 and R4, or proximity to noisy components RUN is brought below its logic threshold. Therefore, in where noise pickup may cause the turn-on or turn-off of the order to completely shut down the IC and reduce the VIN IC to be intermittent. In these cases, a small filter capaci- current to 10nA (typical), it is necessary to ensure that tor can be added across R4 to ensure proper operation. RUN is brought below its worst case low logic threshold of 0.5V. RUN is a high voltage input and can be tied directly PGOOD Comparator to V to continuously enable the IC when the input supply IN The LTC3129-1 provides an open-drain PGOOD output that is present. Also note that RUN can be driven above V IN pulls low if V falls more than 7.5% (typical) below its OUT or V as long as it stays within the operating range of OUT programmed value. When V rises to within 5% (typical) OUT the IC (up to 15V). of its programmed value, the internal PGOOD pull-down With the addition of an optional resistor divider as shown will turn off and PGOOD will go high if an external pull- in Figure 2, the RUN pin can be used to establish a user- up resistor has been provided. An internal filter prevents programmable turn-on and turn-off threshold. This feature nuisance trips of PGOOD due to short transients on V . OUT can be utilized to minimize battery drain below a certain Note that PGOOD can be pulled up to any voltage, as long input voltage, or to operate the converter in a hiccup mode as the absolute maximum rating of 18V is not exceeded, from very low current sources. and as long as the maximum sink current rating is not exceeded when PGOOD is low. Note that PGOOD will also be driven low if V is below its UVLO threshold or CC 31291fc 15 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion if the part is in shutdown (RUN below its logic threshold) The MPPC feature provides capabilities to the LTC3129-1 while V is being held up (or back-driven). PGOOD is that can ease the design of intrinsically safe power sup- CC not affected by V UVLO or the accurate RUN threshold. plies. For an example of an application that must operate IN from a supply with intentional series resistance, refer to In cases where V is not being back-driven in shutdown, CC the application example on the bottom of page 25. PGOOD will not be held low indefinitely. The internal PGOOD pull-down will be disabled as the V voltage decays below Note that external compensation should not be required CC approximately 1V. for MPPC loop stability if the input filter capacitor, C , is IN at least 22µF. See Typical Applications for an example of Maximum Power-Point Control (MPPC) external compensation that can be added in applications where C must be less than the recommended minimum The MPPC input of the LTC3129-1 can be used with an IN value. optional external voltage divider to dynamically adjust the commanded inductor current in order to maintain The divider resistor values can be in the megohm range to a minimum input voltage when using high resistance minimize the input current in very low power applications. sources, such as photovoltaic panels, so as to maximize However, stray capacitance and noise pickup on the MPPC input power transfer and prevent VIN from dropping too pin must also be minimized. low under load. Referring to Figure 3, the MPPC pin is The MPPC pin controls the converter in a linear fashion internally connected to the noninverting input of a g m when using sources that can provide a minimum of 5mA amplifier, whose inverting input is connected to the 1.175V to 10mA of continuous input current. For operation from reference. If the voltage at MPPC, using the external volt- weaker input sources, refer to the Application Information age divider, falls below the reference voltage, the output of section to see how the programmable RUN pin can be used the amplifier pulls the internal V node low. This reduces C to control the converter in a hysteretic manner to provide the commanded average inductor current so as to reduce an effective MPPC function for sources that can provide the input current and regulate V to the programmed IN as little as 5µA or less. minimum voltage, as given by: If the MPPC function is not required, the MPPC pin should V = 1.175V • (1 + R5/R6) IN(MPPC) be tied to V . CC V Programming Pins VIN OUT The LTC3129-1 has a precision internal voltage divider on RS *CIN R5 VIN LTC3129-1 V , eliminating the need for high-value external feedback OUT + MPPC + resistors. This not only eliminates two external compo- VSOURCE R6 – nents, it minimizes no-load quiescent current by using very – 1.175V + * CLEINA SSTH O2U2µLFD F BOER AT – VCCURRENT MPPC APPLICATIONS COMMAND VOLTAGE ERROR AMP 31291 F03 Figure 3. MPPC Amplifier with External Resistor Divider 31291fc 16 For more information www.linear.com/LTC3129-1

LTC3129-1 operaTion high resistance values that would not be practical due to the dissipation of the IC. As described elsewhere in this data effects of noise and board leakages that would cause V sheet, bootstrapping of the V for 5V output applications OUT CC regulation errors. The tap point on this divider is digitally can essentially eliminate the V power dissipation term CC selected by using the VS1, VS2 and VS3 pins to program and significantly improve efficiency. As a result, careful one of eight fixed output voltages. The VS pins should be consideration must be given to the thermal environment grounded or connected to V to select the desired output of the IC in order to provide a means to remove heat from CC voltage, according to the following table. The VS1, VS2 the IC and ensure that the LTC3129-1 is able to provide and VS3 pins can also be driven by external logic signals its full rated output current. Specifically, the exposed die as long as the absolute maximum voltage ratings are not attach pad of both the QFN and MSE packages must be exceeded. Note however that driving any of the voltage soldered to a copper layer on the PCB to maximize the select pins high to a voltage less than the V operating conduction of heat out of the IC package. This can be ac- CC voltage will result in increased quiescent current. Also complished by utilizing multiple vias from the die attach note that if the VS3 pin is driven above V , an external pad connection underneath the IC package to other PCB CC 1M resistor should be added in series. For other output layer(s) containing a large copper plane. A typical board voltages, refer to the LTC3129 which has a feedback pin, layout incorporating these concepts is shown in Figure 4. allowing any output voltage from 1.4V to 15.75V. If the IC die temperature exceeds approximately 180°C, V Program Settings for the LTC3129-1 overtemperature shutdown will be invoked and all switching OUT will be inhibited. The part will remain disabled until the die VS3 PIN VS2 PIN VS1 PIN V OUT temperature cools by approximately 10°C. The soft-start 0 0 0 2.5V circuit is re-initialized in over temperature shutdown to 0 0 V 3.3V CC provide a smooth recovery when the IC die temperature 0 V 0 4.1V CC cools enough to resume operation. 0 V V 5.0V CC CC V 0 0 6.9V CC VCC 0 VCC 8.2V GND VIN V V 0 12V CC CC VCC VCC VCC 15V CIN VCC Note that in shutdown, or if V is below its UVLO thresh- CC CBST1 old, the internal voltage divider on V is automatically OUT L disconnected to eliminate any current draw on V . OUT CBST2 Thermal Considerations The power switches of the LTC3129-1 are designed to op- COUT erate continuously with currents up to the internal current 31291 F04 limit thresholds. However, when operating at high current GND VOUT levels, there may be significant heat generated within the IC. In addition, the V regulator can also generate wasted CC Figure 4. Typical 2-Layer PC Board Layout (MSE Package) heat when V is very high, adding to the total power IN 31291fc 17 For more information www.linear.com/LTC3129-1

LTC3129-1 applicaTions inForMaTion A standard application circuit for the LTC3129-1 is shown only applications can use the larger inductor values as on the front page of this data sheet. The appropriate selec- they are unaffected by the RHP zero, while mostly boost tion of external components is dependent upon the required applications generally require inductance on the low end performance of the IC in each particular application given of this range depending on how large the step-up ratio is. considerations and trade-offs such as PCB area, input Regardless of inductor value, the saturation current rating and output voltage range, output voltage ripple, transient should be selected such that it is greater than the worst response, required efficiency, thermal considerations and case average inductor current plus half of the ripple cur- cost. This section of the data sheet provides some basic rent. The peak-to-peak inductor current ripple for each guidelines and considerations to aid in the selection of operational mode can be calculated from the following external components and the design of the applications formula, where f is the switching frequency (1.2MHz), L circuit, as well as more application circuit examples. is the inductance in µH and t is the switch pin mini- LOW mum low time in µs. The switch pin minimum low time V Capacitor Selection CC is typically 0.09µs. The V output of the LTC3129-1 is generated from V CC IN V ⎛V –V ⎞⎛1 ⎞ by a low dropout linear regulator. The VCC regulator has ΔIL(P−P)(BUCK)= OUT⎜ IN OUT⎟⎜ –tLOW⎟ A been designed for stable operation with a wide range L ⎝ VIN ⎠⎝f ⎠ of output capacitors. For most applications, a low ESR V ⎛V –V ⎞⎛1 ⎞ capacitor of at least 2.2µF should be used. The capacitor ΔI = IN⎜ OUT IN⎟⎜ –t ⎟ A L(P−P)(BOOST) LOW should be located as close to the VCC pin as possible and L ⎝ VOUT ⎠⎝f ⎠ connected to the V pin and ground through the shortest CC It should be noted that the worst-case peak-to-peak in- traces possible. V is the regulator output and is also the CC ductor ripple current occurs when the duty cycle in buck internal supply pin for the LTC3129-1 control circuitry as mode is minimum (highest V ) and in boost mode when well as the gate drivers and boost rail charging diodes. IN the duty cycle is 50% (V = 2 • V ). As an example, if The V pin is not intended to supply current to other OUT IN CC V (minimum) = 2.5V and V (maximum) = 15V, V external circuitry. IN IN OUT = 5V and L = 10µH, the peak-to-peak inductor ripples at Inductor Selection the voltage extremes (15V VIN for buck and 2.5V VIN for boost) are: The choice of inductor used in LTC3129-1 application cir- cuits influences the maximum deliverable output current, BUCK = 248mA peak-to-peak the converter bandwidth, the magnitude of the inductor BOOST = 93mA peak-to-peak current ripple and the overall converter efficiency. The One half of this inductor ripple current must be added to inductor must have a low DC series resistance, when compared to the internal switch resistance, or output the highest expected average inductor current in order to current capability and efficiency will be compromised. select the proper saturation current rating for the inductor. Larger inductor values reduce inductor current ripple To avoid the possibility of inductor saturation during load but may not increase output current capability as is the transients, an inductor with a saturation current rating of case with peak current mode control as described in the at least 600mA is recommended for all applications. Maximum Output Current section. Larger value inductors also tend to have a higher DC series resistance for a given In addition to its influence on power conversion efficiency, case size, which will have a negative impact on efficiency. the inductor DC resistance can also impact the maximum Larger values of inductance will also lower the right half output current capability of the buck-boost converter plane (RHP) zero frequency when operating in boost mode, particularly at low input voltages. In buck mode, the which can compromise loop stability. Nearly all LTC3129-1 output current of the buck-boost converter is primarily application circuits deliver the best performance with limited by the inductor current reaching the average cur- an inductor value between 3.3µH and 10µH. Buck mode rent limit threshold. However, in boost mode, especially 31291fc 18 For more information www.linear.com/LTC3129-1

LTC3129-1 applicaTions inForMaTion at large step-up ratios, the output current capability can Recommended inductor values for different operating also be limited by the total resistive losses in the power voltage ranges are given in Table 3. These values were stage. These losses include, switch resistances, inductor chosen to minimize inductor size while maintaining an DC resistance and PCB trace resistance. Avoid inductors acceptable amount of inductor ripple current for a given with a high DC resistance (DCR) as they can degrade the V and V range. IN OUT maximum output current capability from what is shown Table 3. Recommended Inductor and Output Capacitor Values in the Typical Performance Characteristics section and V AND V RANGE RECOMMENDED MAXIMUM RECOMMENDED from the Typical Application circuits. IN OUT INDUCTOR TOTAL OUTPUT CAPACITOR VALUES VALUE FOR PWM MODE As a guideline, the inductor DCR should be significantly OPERATION AT LIGHT LOAD less than the typical power switch resistance of 750mΩ (<15mA, PWM PIN HIGH) each. The only exceptions are applications that have a V and V Both < 4.5V 3.3µH to 4.7µH 10µF IN OUT maximum output current requirement much less than V and V Both < 8V 4.7µH to 6.8µH 10µF IN OUT what the LTC3129-1 is capable of delivering. Generally V and V Both < 11V 6.8µH to 8.2µH 10µF IN OUT speaking, inductors with a DCR in the range of 0.15Ω to V and V Up to 15V 8.2µH to 10µH 10µF IN OUT 0.3Ω are recommended. Lower values of DCR will improve the efficiency at the expense of size, while higher DCR Due to the fixed, internal loop compensation and feedback values will reduce efficiency (typically by a few percent) divider provided by the LTC3129-1, there are limitations to while allowing the use of a physically smaller inductor. the maximum recommended total output capacitor value in applications that must operate in PWM mode at light load Different inductor core materials and styles have an impact (PWM pin pulled high with minimum load currents less on the size and price of an inductor at any given current rating. Shielded construction is generally preferred as it than ~15mA). In these applications, a maximum output minimizes the chances of interference with other circuitry. capacitor value, shown in Table 3, is recommended. For The choice of inductor style depends upon the price, sizing, applications that must operate in PWM mode at light load and EMI requirements of a particular application. Table 2 with higher values of output capacitance, the LTC3129 is provides a wide sampling of inductors that are well suited recommended. Its external feedback pin allows the use to many LTC3129-1 applications. of additional feedforward compensation for improved light-load stability under these conditions. Table 2. Recommended Inductors Note that for applications where Burst Mode operation VENDOR PART is enabled (PWM pin grounded), the output capacitor Coilcraft EPL2014, EPL3012, EPL3015, XFL3012 www.coilcraft.com LPS3015, LPS3314 value can be increased without limitation regardless of Coiltronics SDH3812, SD3814 the minimum load current or inductor value. www.cooperindustries.com SD3114, SD3118 Murata LQH3NP Output Capacitor Selection www.murata.com LQH32P LQH44P A low effective series resistance (ESR) output capacitor Sumida CDRH2D16, CDRH2D18 of 4.7µF minimum should be connected at the output of www.sumida.com CDRH3D14, CDRH3D16 the buck-boost converter in order to minimize output volt- Taiyo-Yuden NR3012T, NR3015T, NRS4012T age ripple. Multilayer ceramic capacitors are an excellent www.t-yuden.com BRC2518 option as they have low ESR and are available in small TDK VLS3012, VLS3015 footprints. The capacitor value should be chosen large www.tdk.com VLF302510MT, VLF302512MT enough to reduce the output voltage ripple to acceptable Toko DB3015C, DB3018C, DB3020C www.tokoam.com DP418C, DP420C, DEM2815C, levels. Neglecting the capacitor’s ESR and ESL (effec- DFE322512C, DFE252012C tive series inductance), the peak-to-peak output voltage Würth WE-TPC 2813, WE-TPC 3816, ripple in PWM mode can be calculated by the following www.we-online.com WE-TPC 2828 31291fc 19 For more information www.linear.com/LTC3129-1

LTC3129-1 applicaTions inForMaTion formula, where f is the frequency in MHz (1.2MHz), C Input Capacitor Selection OUT is the capacitance in µF, t is the switch pin minimum LOW The V pin carries the full inductor current and provides IN low time in µs (0.09µs typical) and I is the output LOAD power to internal control circuits in the IC. To minimize current in amperes. input voltage ripple and ensure proper operation of the IC, I t a low ESR bypass capacitor with a value of at least 4.7µF LOAD LOW ΔV = V P−P(BUCK) should be located as close to the V pin as possible. The C IN OUT traces connecting this capacitor to V and the ground IN ILOAD ⎛VOUT –VIN+tLOWfVIN⎞ plane should be made as short as possible. ΔV = ⎜ ⎟ V P−P(BOOST) fCOUT⎝ VOUT ⎠ When powered through long leads or from a power source with significant resistance, a larger value bulk input ca- Examining the previous equations reveals that the output pacitor may be required and is generally recommended. voltage ripple increases with load current and is gener- In such applications, a 47µF to 100µF low-ESR electrolytic ally higher in boost mode than in buck mode. Note that capacitor in parallel with a 1µF ceramic capacitor generally these equations only take into account the voltage ripple yields a high performance, low cost solution. that occurs from the inductor current to the output being discontinuous. They provide a good approximation to the Note that applications using the MPPC feature should ripple at any significant load current but underestimate the use a minimum CIN of 22µF. Larger values can be used output voltage ripple at very light loads where the output without limitation. voltage ripple is dominated by the inductor current ripple. Recommended Input and Output Capacitor Types In addition to the output voltage ripple generated across The capacitors used to filter the input and output of the the output capacitance, there is also output voltage ripple LTC3129-1 must have low ESR and must be rated to handle produced across the internal resistance of the output the AC currents generated by the switching converter. capacitor. The ESR-generated output voltage ripple is This is important to maintain proper functioning of the proportional to the series resistance of the output capacitor IC and to reduce output voltage ripple. There are many and is given by the following expressions where R is ESR capacitor types that are well suited to these applications the series resistance of the output capacitor and all other including multilayer ceramic, low ESR tantalum, OS-CON terms as previously defined. and POSCAP technologies. In addition, there are certain I R LOAD ESR types of electrolytic capacitors such as solid aluminum ΔV = ≅I R V P−P(BUCK) 1–t f LOAD ESR organic polymer capacitors that are designed for low ESR LOW and high AC currents and these are also well suited to I R V LOAD ESR OUT ΔV = some LTC3129-1 applications. The choice of capacitor P−P(BOOST) ( ) V 1–t f technology is primarily dictated by a trade-off between IN LOW size, leakage current and cost. In backup power applica- ⎛V ⎞ ≅I R ⎜ OUT⎟V tions, the input or output capacitor might be a super or LOAD ESR ⎝ VIN ⎠ ultra capacitor with a capacitance value measuring in the farad range. The selection criteria in these applications In most LTC3129-1 applications, an output capacitor be- are generally similar except that voltage ripple is generally tween 10µF and 22µF will work well. To minimize output not a concern. Some capacitors exhibit a high DC leak- ripple in Burst Mode operation, values of 22µF operation age current which may preclude their consideration for or larger are recommended. applications that require a very low quiescent current in Burst Mode operation. Note that ultra capacitors may have 31291fc 20 For more information www.linear.com/LTC3129-1

LTC3129-1 applicaTions inForMaTion a rather high ESR, therefore a 4.7µF (minimum) ceramic Although the converter will be operating in bursts, it is capacitor is recommended in parallel, close to the IC pins. enough to charge an output capacitor to power low duty cycle loads, such as wireless sensor applications, or to Ceramic capacitors are often utilized in switching con- trickle charge a battery. In addition, note that the input verter applications due to their small size, low ESR and voltage will be cycling (with a small ripple as set by the low leakage currents. However, many ceramic capacitors RUN hysteresis) about a fixed voltage, as determined by intended for power applications experience a significant the divider. This allows the high impedance source to loss in capacitance from their rated value as the DC bias operate at the programmed optimal voltage for maximum voltage on the capacitor increases. It is not uncommon for power transfer. a small surface mount capacitor to lose more than 50% of its rated capacitance when operated at even half of its When using high value divider resistors (in the MΩ range) maximum rated voltage. This effect is generally reduced to minimize current draw on VIN, a small noise filter ca- as the case size is increased for the same nominal value pacitor may be necessary across the lower divider resis- capacitor. As a result, it is often necessary to use a larger tor to prevent noise from erroneously tripping the RUN value capacitance or a higher voltage rated capacitor than comparator. The capacitor value should be minimized would ordinarily be required to actually realize the intended so as not to introduce a time delay long enough for the capacitance at the operating voltage of the application. X5R input voltage to drop significantly below the desired VIN and X7R dielectric types are recommended as they exhibit threshold before the converter is turned off. Note that the best performance over the wide operating range and larger VIN decoupling capacitor values will minimize this temperature of the LTC3129-1. To verify that the intended effect by providing more holdup time on VIN. capacitance is achieved in the application circuit, be sure Programming the MPPC Voltage to consult the capacitor vendor’s curve of capacitance versus DC bias voltage. As discussed in the previous section, the LTC3129-1 in- cludes an MPPC function to optimize performance when Using the Programmable RUN Function to Operate operating from voltage sources with relatively high source from Extremely Weak Input Sources resistance. Using an external voltage divider from V , the IN Another application of the programmable RUN pin is that MPPC function takes control of the average inductor current it can be used to operate the converter in a hiccup mode when necessary to maintain a minimum input voltage, as from extremely low current sources. This allows opera- programmed by the user. Referring to Figure 3: tion from sources that can only generate microamps of V = 1.175V • (1 + R5/R6) IN(MPPC) output current, and would be far too weak to sustain normal steady-state operation, even with the use of the This is useful for such applications as photovoltaic pow- MPPC pin. Because the LTC3129-1 draws only 1.9µA ered converters, since the maximum power transfer point typical from V until it is enabled, the RUN pin can be occurs when the photovoltaic panel is operated at about IN programmed to keep the IC disabled until V reaches the 75% of its open-circuit voltage. For example, when operat- IN programmed voltage level. In this manner, the input source ing from a photovoltaic panel with an open-circuit voltage can trickle-charge an input storage capacitor, even if it of 5V, the maximum power transfer point will be when can only supply microamps of current, until V reaches the panel is loaded such that its output voltage is about IN the turn-on threshold set by the RUN pin divider. The 3.75V. Choosing values of 2MΩ for R5 and 909k for R6 converter will then be enabled, using the stored charge will program the MPPC function to regulate the maximum in the input capacitor, until V drops below the turn-off input current so as to maintain V at a minimum of 3.74V IN IN threshold, at which point the converter will turn off and (typical). Note that if the panel can provide more power the process will repeat. than the LTC3129-1 can draw, the input voltage will rise above the programmed MPPC point. This is fine as long This approach allows the converter to run from weak as the input voltage doesn't exceed 15V. sources such as thin-film solar cells using indoor lighting. 31291fc 21 For more information www.linear.com/LTC3129-1

LTC3129-1 applicaTions inForMaTion For weak input sources with very high resistance (hun- Powering V in this manner is referred to as bootstrap- CC dreds of Ohms or more), the LTC3129-1 may still draw ping. This can be done by connecting a Schottky diode more current than the source can provide, causing V to (such as a BAT54) from V to V as shown in Figure 5. IN OUT CC drop below the UVLO threshold. For these applications, it With the bootstrap diode installed, the gate driver currents is recommended that the programmable RUN feature be are supplied by the buck-boost converter at high efficiency used, as described in the previous section. rather than through the internal linear regulator. The in- ternal linear regulator contains reverse blocking circuitry MPPC Compensation and Gain that allows V to be driven above its nominal regulation CC level with only a very slight amount of reverse current. When using MPPC, there are a number of variables that Please note that the bootstrapping supply (either V or affect the gain and phase of the input voltage control OUT a separate regulator) must be limited to less than 5.7V so loop. Primarily these are the input capacitance, the MPPC as not to exceed the maximum V voltage of 5.5V after divider ratio and the V source resistance (or current). To CC IN the diode drop. simplify the design of the application circuit, the MPPC control loop in the LTC3129 is designed with a relatively By maintaining V above its UVLO threshold, bootstrap- CC low gain, such that external MPPC loop compensation is ping, even to a 3.3V output, also allows operation down generally not required when using a VIN capacitor value to the VIN UVLO threshold of 1.8V (typical). of at least 22µF. The gain from the MPPC pin to the in- ternal VC control voltage is about 12, so a drop of 50mV on the MPPC pin (below the 1.175V MPPC threshold), VOUT VOUT corresponds to a 600mV drop on the internal VC voltage, LTC3129-1 COUT BAT54 which reduces the average inductor current all the way to zero. Therefore, the programmed input MPPC voltage VCC will be maintained within about 4% over the load range. 2.2µF Note that if large-value V capacitors are used (which may 31291 F05 IN have a relatively high ESR) a small ceramic capacitor of Figure 5. Example of V Bootstrap at least 4.7µF should be placed in parallel across the V CC IN input, near the V pin of the IC. IN Sources of Small Photovoltaic Panels Bootstrapping the V Regulator CC A list of companies that manufacture small solar panels The high and low side gate drivers are powered through (sometimes referred to as modules or solar cell arrays) the V rail, which is generated from the input voltage, V , suitable for use with the LTC3129-1 is provided in Table 4. CC IN through an internal linear regulator. In some applications, Table 4. Small Photovoltaic Panel Manufacturers especially at high input voltages, the power dissipation Sanyo http://panasonic.net/energy/amorton/en/ in the linear regulator can become a major contributor to PowerFilm http://www.powerfilmsolar.com/ thermal heating of the IC and overall efficiency. The Typical Performance Characteristics section provides data on the IXYS http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx Corporation V current and resulting power loss versus V and V . CC IN OUT G24 http://www.g24i.com/ A significant performance advantage can be attained in high Innovations V applications where converter output voltage (V ) is IN OUT programmed to 5V, if V is used to power the V rail. OUT CC 31291fc 22 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical applicaTions Low Noise, Fixed Frequency, Wide V Range 12V Converter IN 22nF 22nF 6.8µH VIN < 12V, IOUT = 30mA VIN > 12V, IOUT = 200mA BST1 SW1 SW2 BST2 2.42V TO 1V5IVN VIN VOUT V12OVUT 10µF LTC3129-1 16V RUN 1M 4.7µF PGOOD PGOOD VCC MPPC PWM VS1 VCC VS2 2.2µF VS3 GND PGND 31291 TA02 3.3V Converter Provides Extremely Long Run Time in Low Drain Applications Using Lithium Thionyl Chloride Battery 22nF 22nF 4.2µH BST1 SW1 SW2 BST2 VIN VIN VOUT V3.O3UVT 47µF 22µF LTC3129-1 RUN 1M VCC MPPC PGOOD PGOOD PWM Li-SoCl2 AA VS1 VCC SAFT LS14500 TADIRAN TL-4903 VS2 2.2µF VS3 GND PGND 31291 TA03 RUN TIME > 100,000 HRS (11.4 YEARS) AT 10µA (33µW) AVERAGE LOAD > 34,000 HRS (3.9 YEARS) AT 50µA (165µW) AVERAGE LOAD 31291fc 23 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical applicaTions 15V Converter Powered from Flexible Solar Panel I vs Light Level (Daylight) OUT 22nF 22nF 10µH 100 VIN VMPPC = 6V VBISNT1 SW1 SW2 BVSOTU2T IOUT = 32mAV1 5OINVU TFULL SUN 1M 10µF LTC3129-1 A) 47µF RUN m (UT 10 O I PowerFilm MPPC PGOOD MPT6-150 SOLAR PWM MODULE VCC VS1 VCC 11.4cm × 15cm VS2 1 243k VS3 GND PGND 2.2µF 10000 100000 1000000 LIGHT LEVEL (Lx) 31291 TA04b 31291 TA04a Hiccup Converter Keeps Li-Ion Battery Charged with Indoor Lighting Average I vs Light Level OUT (Indoors) 22nF 22nF 3.3µH 1000 VIN UVLO = 3.5V VBISNT1 SW1 SW2 BVSOTU2T 4V.O1UVT + 4.42M 4.7µF LTC3129-1 4.7µF Li-Ion 470µF A) µ 6.3V RUN (UT100 O I PV PANEL VCC MPPC PGOOD SANYO AM-1815 PWM 4.9cm × 5.8cm 10pF VS1 VCC 2.37M VS2 10 2.2µF 100 1000 10000 VS3 GND PGND LIGHT LEVEL (Lx) 31291 TA05b 31291 TA05a 31291fc 24 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical applicaTions 5V Converter Operates from Two to Eight AA or AAA Cells Using Bootstrap Diode to Increase Efficiency at High V and Extend Operation at Low V IN IN 22nF 22nF 8.2µH VIN < 5V, IOUT = 100mA VIN > 5V, IOUT = 200mA BST1 SW1 SW2 BST2 1.92V TO 1V5IVN VIN VOUT V5VOUT 22µF LTC3129-1 AFTER STARTUP RUN BAT54 TWO TO EIGHT VCC MPPC PGOOD AA OR AAA BATTERIES PWM 10µF VS1 VCC VS2 2.2µF VS3 GND PGND 31291 TA06 3.3V Converter Uses MPPC Function to Work with High Resistance Battery Pack 22nF 22nF 3.3µH VIN VMPPC = 2.9V VBISNT1 SW1 SW2 BVSOTU2T IOUT = 100mV3A.O3UVT 10µF LTC3129-1 10Ω 10µF 1.5M RUN MPPC PGOOD 1.5V PWM 1.5V R15C0k VCC VS1 VCC 1.5V C33CpF 1M VVSS23 GND PGND 2.2µF 31291 TA07 NOTE: RC AND CC HAVE BEEN ADDED FOR IMPROVED MPPC LOOP STABILITY WHEN USING AN INPUT CAPACITOR VALUE LESS THAN THE RECOMMENDED MINIMUM OF 22µF 31291fc 25 For more information www.linear.com/LTC3129-1

LTC3129-1 Typical applicaTions Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications 22nF 22nF 3.3µH FDC6312P DUAL PMOS VOUT 3V TO 3.3V VIN UVLO = 3.7V BST1 SW1 SW2 BST2 3.30V D1 S1 S2 D2 2.2µF VIN VOUT 22µF 4.7µF 4.99M LTC3129-1 2.43M G1 G2 RUN CR2032 PV PANEL + VOUT 3V COIN CELL SANYO AM-1815 470µF VCC MPPC OR 6.3V PowerFilm SP4.2-37 PWM PGOOD 2.43M VS1 BAT54 VS2 10pF VCC 74LVC2G04 VS3 GND PGND 2.2µF 31291 TA09 Percentage of Added Battery Life vs Light Level and Load (PowerFilm SP4.2-37, 30sq cm Panel) 1000 %) E ( 100 LIF Y R E ATT B ED 10 DD AVERAGE LOAD = 165µW A AVERAGE LOAD = 330µW AVERAGE LOAD = 660µW AVERAGE LOAD = 1650µW AVERAGE LOAD = 3300µW 1 100 1,000 10,000 LIGHT LEVEL (Lx) 31291 TA09b 31291fc 26 For more information www.linear.com/LTC3129-1

LTC3129-1 package DescripTion Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings. UD Package 16-Lead Plastic QFN (3mm × 3mm) UD Package (Reference LTC DWG # 05-08-1700 Rev A) 16-Lead Plastic QFN (3mm × 3mm) (RefereEnxcpeo sLeTdC P DaWd VGa #ri a0t5io-0n8 A-A1700 Rev A) Exposed Pad Variation AA 0.70 ±0.05 3.50 ±0.05 1.65 ±0.05 2.10 ±0.05 (4 SIDES) PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS BOTTOM VIEW—EXPOSED PAD R = 0.115 PIN 1 NOTCH R = 0.20 TYP 3.00 ±0.10 0.75 ±0.05 TYP OR 0.25 × 45° CHAMFER (4 SIDES) 15 16 PIN 1 0.40 ±0.10 TOP MARK (NOTE 6) 1 1.65 ±0.10 2 (4-SIDES) (UD16 VAR A) QFN 1207 REV A 0.200 REF 0.25 ±0.05 0.00 – 0.05 0.50 BSC NOTE: 1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 31291fc 27 For more information www.linear.com/LTC3129-1

LTC3129-1 package DescripTion Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings. MSE Package 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.112 ±.004) (.035 ±.005) 1 8 0.35 REF 5.10 1.651 ±0.102 3.20 – 3.45 1.651 ±0.102 (.M20IN1) (.065 ±.004) (.126 – .136) (.065 ±.004) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 16 9 0.305 ±0.038 0.50 NO MEASUREMENT PURPOSE (.0120 ±.0015) (.0197) 4.039 ±0.102 TYP BSC (.159 ±.004) (NOTE 3) 0.280 ±0.076 RECOMMENDED SOLDER PAD LAYOUT 16151413121110 9 (.011 ±.003) REF DETAIL “A” 0.254 (.010) 3.00 ±0.102 0° – 6° TYP 4.90 ±0.152 (.118 ±.004) (.193 ±.006) GAUGE PLANE (NOTE 4) 0.53 ±0.152 (.021 ±.006) 1234567 8 DETAIL “A” 1.10 0.86 0.18 (.043) (.034) (.007) MAX REF SEATING PLANE 0.17 – 0.27 0.1016 ±0.0508 (.007 – .011) (.004 ±.002) TYP 0.50 NOTE: (.0197) MSOP (MSE16) 0213 REV F 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 31291fc 28 For more information www.linear.com/LTC3129-1

LTC3129-1 revision hisTory REV DATE DESCRIPTION PAGE NUMBER A 5/14 Clarified V Leakage to V if V > V : from –7µA to –27µA 4 CC IN CC IN B 10/14 Clarified PGOOD Pin Description 9 Clarified Operation Paragraph 16 C 10/15 Changed MAX V Current Limit 4 CC Modified MPPC section 16 Modified Table 4 22 31291fc Information furnished by Linear Technology Corporation is believed to be accurate and reliable. 29 However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnFeoctri omn oofr iets i cnifrocurimts aatsi odnes wcrwibewd. lhinereeainr. wcoillm no/Lt TinCfr3in1g2e9 o-n1 existing patent rights.

LTC3129-1 Typical applicaTion TEG Powered Converter Operates from a 10°C Temperature Differential and Provides 3.3V at 25mA for 50ms Every 15 Seconds for a Wireless Sensor COILCRAFT LPR6235-123QML 33nF 1:50 + C1A VSTORE + 220µF • • 1nF C1B 470µF 6.3V C2A MTEAGR MLOOWUN NTLE1D0 2T5OT 330k C2B VOUT2 LTC3109 A HEAT SINK WITH LESS THAN 15°C/W SWA THERMAL RESISTANCE SWB VOUT2_EN VINA VINB VOUT VAUX VS2 PGOOD VS1 VLDO VAUX VAUX 1µF 22nF 22nF 4.7µH BST1 SW1 SW2 BST2 1µF VIN VOUT V3.O3UVT 3.01M LTC3129-1 10µF 1M RUN 10pF 1M VCC PGOOD PGOOD MPPC BAT54 PWM 1N4148 VS1 VCC 2.2µF VS2 1M VS3 GND PGND 31291 TA08 relaTeD parTs PART NUMBER DESCRIPTION COMMENTS LTC3103 15V, 300mA Synchronous Step-Down DC/DC Converter with V = 2.2V, V = 15V, V = 0.8V, I = 1.8µA, IN(MIN) IN(MAX) OUT(MIN) Q Ultralow Quiescent Current I <1µA, 3mm × 3mm DFN-10, MSOP-10 Packages SD LTC3104 15V, 300mA Synchronous Step-Down DC/DC Converter with V = 2.2V, V = 15V, V = 0.8V, I = 2.8µA, IN(MIN) IN(MAX) OUT(MIN) Q Ultralow Quiescent Current and 10mA LDO I <1µA, 4mm × 3mm DFN-14, MSOP-16 Packages SD LTC3105 400mA Step-up Converter with MPPC and 250mV Start-Up V = 0.2V, V = 5V, V = 0 5.25V , I = 22µA, IN(MIN) IN(MAX) OUT(MIN) MAX Q I <1µA, 3mm × 3mm DFN-10/MSOP-12 Packages SD LTC3112 15V, 2.5A, 750kHz Monolithic Synch Buck/Boost V = 2.7V, V = 15V, V = 2.7V to 14V, I = 50µA, IN(MIN) IN(MAX) OUT(MIN) Q I <1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages SD LTC3115-1 40V, 2A, 2MHz Monolithic Synch Buck/Boost V = 2.7V, V = 40V, V = 2.7V to 40V, I = 50µA, IN(MIN) IN(MAX) OUT(MIN) Q I <1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages SD LTC3531 5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost V = 1.8V, V = 5.5V, V = 2V to 5V, I = 16µA, IN(MIN) IN(MAX) OUT(MIN) Q I <1µA, 3mm × 3mm DFN-8 and ThinSOT Packages SD LTC3388-1/ 20V, 50mA High Efficiency Nano Power Step-Down Regulator V = 2.7V, V =20V, V = Fixed 1.1V to 5.5V, IN(MIN) IN(MAX) OUT(MIN) LTC3388-3 I = 720nA, I = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages Q SD LTC3108/ Ultralow Voltage Step-Up Converter and Power Manager V = 0.02V, V = 1V, V = Fixed 2.35V to 5V, IN(MIN) IN(MAX) OUT(MIN) LTC3108-1 I = 6µA, I <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages Q SD LTC3109 Auto-Polarity, Ultralow Voltage Step-Up Converter and Power V = 0.03V, V = 1V, V = Fixed 2.35V to 5V, IN(MIN) IN(MAX) OUT(MIN) Manager I = 7µA, I <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages Q SD LTC3588-1 Piezo Electric Energy Harvesting Power Supply V = 2.7V, V = 20V, V = Fixed 1.8V to 3.6V, IN(MIN) IN(MAX) OUT(MIN) I = 950nA, I 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages Q SD LTC4070 Li-Ion/Polymer Low Current Shunt Battery Charger System V = 450nA to 50mA, V + 4.0V, 4.1V, 4.2V, I = 300nA, IN(MIN) FLOAT Q 2mm × 3mm DFN-8, MSOP-8 Packages 31291fc 30 Linear Technology Corporation LT 1015 REV C • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95F0or3 m5-o7r4e 1in7formation www.linear.com/LTC3129-1 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/3129-1  LINEAR TECHNOLOGY CORPORATION 2013