LTC1622 Low Input Voltage Current Mode Step-Down DC/DC Controller FEATURES DESCRIPTIOU n High Efficiency The LTC®1622 is a constant frequency current mode step- n Constant Frequency 550kHz Operation down DC/DC controller providing excellent AC and DC load n V Range: 2V to 10V and line regulation. The device incorporates an accurate IN n Multiampere Output Currents undervoltage feature that shuts the LTC1622 down when n OPTI-LOOPTM Compensation Minimizes C the input voltage falls below 2V. OUT n Selectable, Burst Mode Operation The LTC1622 boasts a – 1.9% output voltage accuracy and n Low Dropout Operation: 100% Duty Cycle consumes only 350m A of quiescent current. For applica- n Synchronizable up to 750kHz tions where efficiency is a prime consideration and the n Current Mode Operation for Excellent Line and Load load current varies from light to heavy, the LTC1622 can Transient Response be configured for Burst ModeTM operation. Burst Mode n Low Quiescent Current: 350m A operation enhances low current efficiency and extends n Shutdown Mode Draws Only 15m A Supply Current battery run time. Burst Mode operation is inhibited during n – 1.9% Reference Accuracy synchronization or when the SYNC/MODE pin is pulled low n Available in 8-Lead MSOP to reduce noise and possible RF interference. APPLICATIOU S High constant operating frequency of 550kHz allows the use of a small inductor. The device can also be synchro- n 1- or 2-Cell Li-Ion Powered Applications nized up to 750kHz for special applications. The high n Cellular Telephones frequency operation and the available 8-lead MSOP pack- nWireless Modems age create a high performance solution in an extremely n Portable Computers small amount of PCB area. n Distributed 3.3V, 2.5V or 1.8V Power Systems To further maximize the life of the battery source, the n Scanners P-channel MOSFET is turned on continuously in dropout n Battery-Powered Equipment (100% duty cycle). In shutdown, the device draws a mere , LTC and LT are registered trademarks of Linear Technology Corporation. 15m A. Burst Mode and OPTI-LOOP are a trademarks of Linear Technology Corporation. TYPICAL APPLICATIOU Efficiency vs Load Current with Burst Mode Operation Enabled VIN 2.5V TO 8.5V 100 8 C1 VIN = 4.2V 2 ITH VINSENSE– 1 R0.203Ω 1100VµF 90 VIN = 3.3V R101C2k230pF LSTYCN1C6/2MP2DORDVE 75 Si3443DV4.L71µH V2.O5UVT NCY (%) 8700 VIN = 8.4V VIN = 6V E D1 R3 1.5A CI 4 RUN/SS GND 6 IR10BQ015 159k + C472µF EFFI 60 470pF VFB R4 6V 3 75k 50 VOUT = 2.5V RSENSE = 0.03Ω C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT L1: MURATA LQN6C-4R7 C2: SANYO POSCAP 6TPA47M R2: DALE WSL-1206 0-03Ω 1622 F01a 40 D1: INTERNATIONAL RECTIFIER IR10BQ015 1 10 100 1000 5000 LOAD CURRENT (mA) Figure 1. High Efficiency Step-Down Converter 1622 F01b 1

LTC1622 ABSOLUTE WMAXIWMUWM RATINUGS (Note 1) Input Supply Voltage (V ).........................–0.3V to 10V Operating Temperature Range IN RUN/SS Voltage .......................................–0.3V to 2.4V Commercial............................................ 0(cid:176) C to 70(cid:176) C SYNC/MODE Voltage................................. –0.3V to V Industrial........................................... –45(cid:176) C to 85(cid:176) C IN SENSE– Voltage .......................................... 2.4V to V Junction Temperature (Note 2).............................125(cid:176) C IN PDRV Peak Output Current (<10m s)......................... 1A Lead Temperature (Soldering, 10 sec)..................300(cid:176) C Storage Ambient Temperature Range... –65(cid:176) C to 150(cid:176) C PACKAGE/ORDER IUNFORWMATIOUN ORDER PART ORDER PART NUMBER TOP VIEW NUMBER SENSE– 1 TOP VIEW 8VIN LTC1622CMS8 SENSE– 1 8 VIN LTC1622CS8 ITH 2 7PDRV ITH 2 7 PDRV LTC1622IS8 VFB 3 6GND VFB 3 6 GND RUN/SS 4 5SYNC/MODE MS8 PART MARKING RUN/SS 4 5 SYNC/MODE S8 PART MARKING MS8 PACKAGE 8-LEAD PLASTIC MSOP S8 PACKAGE TJMAX = 125(cid:176)C, q JA = 250(cid:176)C/W LTDB 8-LEAD PLASTIC SO 1622 TJMAX = 125(cid:176)C, q JA = 150(cid:176)C/W 1622I Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The l denotes specifications which apply over the full operating temperature range, otherwise specifications are at T = 25(cid:176) C. V = 4.2V A IN SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS I Feedback Current (Note 3) V = 0.8V 10 70 nA VFB FB V Regulated Feedback Voltage (Note 3) Commercial Grade l 0.785 0.8 0.815 V FB (Note 3) Industrial Grade l 0.780 0.8 0.820 V V Output Overvoltage Lockout Referenced to Nominal V 4 16 25 % OVL OUT D V Reference Voltage Line Regulation V = 4.2V to 8.5V (Note 3) 0.04 0.08 %/V OSENSE IN V Output Voltage Load Regulation Measured in Servo Loop; V = 0.2V to 0.625V 0.3 0.5 % LOADREG ITH Measured in Servo Loop; V = 0.9V to 0.625V –0.3 –0.5 % ITH I Input DC Supply Current (Note 4) S Burst Mode Inhibited V = 2.3V 450 m A IN Sleep Mode V = 0V, V = 2.4V 350 400 m A ITH SYNC/MODE Shutdown V = 0V 15 30 m A RUN/SS Shutdown V = 0V, V = V – 0.1V 4 10 m A RUN/SS IN UVLO V RUN/SS Threshold Commercial Grade l 0.4 0.7 0.9 V RUN/SS Industrial Grade l 0.3 0.7 1.0 V I Soft-Start Current Source V = 0V 1 2.5 5 m A RUN/SS RUN/SS f Oscillator Frequency V = 0.8V 475 550 625 kHz OSC FB V = 0V 75 110 140 kHz FB V SYNC/MODE Threshold V Ramping Down 1 1.5 V SYNC/MODE SYNC/MODE V Undervoltage Lockout V Ramping Down l 1.55 1.92 2.3 V UVLO IN V Ramping Up 1.97 2.36 V IN 2

LTC1622 ELECTRICAL CHARACTERISTICS The l denotes specifications which apply over the full operating temperature range, otherwise specifications are at T = 25(cid:176) C. V = 4.2V A IN SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS PDRV t Gate Drive Rise Time C = 3000pF 80 140 ns r LOAD PDRV t Gate Drive Fall Time C = 3000pF 100 140 ns f LOAD D V Maximum Current Sense Voltage l 80 110 140 mV SENSE(MAX) Note 1: Absolute Maximum Ratings are those values beyond which the life Note 3: The LTC1622 is tested in a feedback loop that servos V to the FB of a device may be impaired. feedback point for the error amplifier (V = 0.8V). ITH Note 2: T is calculated from the ambient temperature T and power Note 4: Dynamic supply current is higher due to the gate charge being J A dissipation P according to the following formula: delivered at the switching frequency. D LTC1622CS8; T = T + (P • 150(cid:176) C/W), J A D LTC1622CMS8; T = T + (P • 250(cid:176) C/W) J A D TYPICAL PERFORWMANUCE CHARACTERISTICS Shutdown Current Maximum Current Sense Voltage vs Supply Voltage RUN/SS Current vs Supply Voltage vs Duty Cycle 45 3.50 110 VIN = 4.2V 40 100 µA) 35 µA)3.00 90 NT ( 30 NT ( mV) UNSYNC RE RE2.50 E ( 80 R 25 R G U U A SHUTDOWN C 211050 SOFT-START C21..0500 TRIP VOLT 765000 5 40 0 1.00 30 2 3 4 5 6 7 8 9 10 2 3 4 5 6 7 8 9 10 20 30 40 50 60 70 80 90 100 SUPPLY VOLTAGE (V) SUPPLY VOLTAGE (V) DUTY CYCLE (%) 1622 G01 1622 G02 1622 G03 Normalized Oscillator Frequency Reference Voltage Undervoltage Lockout Voltage vs Temperature vs Temperature vs Temperature 10.0 0.810 2.10 VIN = 4.2V VIN = 4.2V V) 7.5 0.805 E ( 2.05 %) AG QUENCY ( 52..05 LTAGE (V)00..870905 OUT VOLT 21..0905 LIZED FRE–2.05 RENCE VO0.790 AGE LOCK 1.90 A E T NORM–5.0 REF0.785 ERVOL 1.85 –7.5 0.780 ND 1.80 U –10.0 0.775 1.75 –55 –35 –15 5 25 45 65 85 105 125 –55 –35 –15 5 25 45 65 85 105 125 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) 1622 G04 1622 G05 1622 G06 3

LTC1622 TYPICAL PERFORWMANUCE CHARACTERISTICS Efficiency vs Load Current for Figure 1 with Burst Mode Load Step Transient Response Load Step Transient Response Operation Defeated Burst Enabled Burst Inhibited 100 VIN = 3.3V 90 VIN = 4.2V V V CIENCY (%) 8700 VIN = 6V 100mV/DI 100mV/DI EFFI 60 VIN = 8.4V 50 VOUT = 2.5V ILOAD = 50mA TO 1.2A ILOAD = 50mA TO 1.2A RSENSE = 0.03Ω VIN = 4.2V VIN = 4.2V 40 1622 G08 1622 G09 1 10 100 1000 LOAD CURRENT (mA) 1622 G07 PIUN FUUNCTIOUNS SENSE– (Pin 1): The Negative Input to the Current Com- SYNC/MODE (Pin 5): This pin performs three functions. parator. Greater than 2V on this pin allows Burst Mode operation at low load currents, while grounding or applying a clock I (Pin 2): Error Amplifier Compensation Point. The TH signal on this pin defeats Burst Mode operation. An current comparator threshold increases with this control external clock between 625kHz and 750kHz applied to this voltage. Nominal voltage range for this pin is 0V to 1.2V. pin forces the LTC1622 to operate at the external clock VFB (Pin 3): Receives the feedback voltage from an exter- frequency. Do not attempt to synchronize below 625kHz. nal resistive divider across the output capacitor. Pin 5 has an internal 1m A pull-up current source. RUN/SS (Pin 4): Combination of Soft-Start and Run GND (Pin 6): Ground Pin. Control Inputs. A capacitor to ground at this pin sets the PDRV (PIN 7): Gate Drive for the External P-Channel ramp time to full output current. The time is approximately MOSFET. This pin swings from 0V to V . 0.45s/m F. Forcing this pin below 0.4V causes all circuitry IN to be shut down. V (Pin 8): Main Supply Pin. Must be closely decoupled IN to ground Pin 6. 4

LTC1622 FUUNCTIOUNAL DIAGRAW VIN VCC Y = “0” ONLY WHEN X IS A CONSTANT “1” BURST DEFEAT OTHERWISE Y = “1” Y 1µA X SLOPE SYNC/ COMP 5 MODE OSC 0.36V 0.3V – VFB 3 + SENSE– 1 8 VIN – EN FREQ SHIFT + – SLEEP 0.8V + VIN VREF EA 0.12V + +ICOMP– – BURST 2.5µA gm = 0.5mΩ VIN SWITCHING 0.8V 2 REFERENCE S LOGIC RUN/SS 4 RUN/ ITH AND VREF SOFT-START R Q BLANKING VIN 0.8V RS1 CIRCUIT PDRV 7 UVLO + TRIP = 1.97V OV 6 VREF + 0.12V – GND SHUTDOWN 1622 BD OPERATIOU (Refer to Functional Diagram) Main Control Loop current source to charge up the soft-start capacitor C . SS When C reaches 0.7V, the main control loop is enabled The LTC1622 is a constant frequency current mode switch- SS with the I voltage clamped at approximately 5% of its ing regulator. During normal operation, the external TH maximum value. As C continues to charge, I is gradu- P-channel power MOSFET is turned on each cycle when SS TH ally released allowing normal operation to resume. the oscillator sets the R latch (R ) and turned off when S S1 the current comparator (I ) resets the latch. The peak Comparator OV guards against transient overshoots COMP inductor current at which I resets the R latch is >16% by turning off the P-channel power MOSFET and COMP S controlled by the voltage on the I pin, which is the output keeping it off until the fault is removed. TH of the error amplifier EA. An external resistive divider connected between V and ground allows EA to receive Burst Mode Operation OUT an output feedback voltage V . When the load current FB The LTC1622 can be enabled to go into Burst Mode increases, it causes a slight decrease in V relative to the FB operation at low load currents simply by leaving the SYNC/ 0.8V reference, which in turn causes the I voltage to TH MODE pin open or connecting it to a voltage of at least 2V. increase until the average inductor current matches the In this mode, the peak current of the inductor is set as if new load current. V = 0.36V (at low duty cycles) even though the voltage ITH The main control loop is shut down by pulling the RUN/SS at the ITH pin is at lower value. If the inductor’s average pin low. Releasing RUN/SS allows an internal 2.5m A current is greater than the load requirement, the voltage at 5

LTC1622 OPERATIOU (Refer to Functional Diagram) the I pin will drop. When the I voltage goes below Short-Circuit Protection TH TH 0.12V, the sleep signal goes high, turning off the external When the output is shorted to ground, the frequency of the MOSFET. The sleep signal goes low when the I voltage TH oscillator will be reduced to about 110kHz. This lower rises above 0.22V and the LTC1622 resumes normal frequency allows the inductor current to safely discharge, operation. The next oscillator cycle will turn the external thereby preventing current runaway. The oscillator’s fre- MOSFET on and the switching cycle repeats. quency will gradually increase to its nominal value when the feedback voltage increases above 0.65V. Note that Frequency Synchronization synchronization is inhibited until the feedback voltage The LTC1622 can be externally driven by a TTL/CMOS goes above 0.3V. compatible clock signal up to 750kHz. Do not synchronize the LTC1622 below its maximum default operating fre- Overvoltage Protection quency of 625kHz as this may cause abnormal operation As a further protection, the overvoltage comparator in the and an undesired frequency spectrum. The LTC1622 is LTC1622 will turn the external MOSFET off when the synchronized to the rising edge of the clock. The external feedback voltage has risen 16% above the reference clock pulse width must be at least 100ns and not more voltage of 0.8V. This comparator has a typical hysteresis than the period minus 200ns. of 35mV. Synchronization is inhibited when the feedback voltage is below 0.3V. This is to prevent inductor current buildup Slope Compensation and Peak Inductor Current under short-circuit conditions. Burst Mode operation is The inductor’s peak current is determined by: deactivated when the LTC1622 is externally driven by a clock. V I = ( ITH ) PK 10 R Dropout Operation SENSE When the input supply voltage decreases towards the when the LTC1622 is operating below 40% duty cycle. output voltage, the rate of change of inductor current However, once the duty cycle exceeds 40%, slope com- during the ON cycle decreases. This reduction means that pensation begins and effectively reduces the peak induc- the P-channel MOSFET will remain on for more than one tor current. The amount of reduction is given by the curves oscillator cycle since the inductor current has not ramped in Figure 2. up to the threshold set by EA. Further reduction in input supply voltage will eventually cause the P-channel MOSFET 110 to be turned on 100%, i.e., DC. The output voltage will then 100 be determined by the input voltage minus the voltage drop 90 across the MOSFET, the sense resistor and the inductor. (%)X) 80 MA 70 Undervoltage Lockout OUT( 60 /IUT 50 IRIPPLE = 0.4IPK Tinop purte vvoelntat goep eleravteilosn, aonf t huen dPe-rcvhoalntangeel MloOckSoFuEtT i bs eilnocwo rspaofe- SF = IO 40 IAARTTIP 55P%%LE DD=UU 0TT.2YYI PCCKYYCCLLEE 30 rated into the LTC1622. When the input supply voltage 20 VIN = 4.2V drops below 2V, the P-channel MOSFET and all circuitry is UNSYNC 10 turned off except the undervoltage block, which draws 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) only several microamperes. 1622 F02 Figure 2. Maximum Output Current vs Duty Cycle 6

LTC1622 APPLICATIOUNS INUFORWMATIOUN The basic LTC1622 application circuit is shown in Figure V . The inductor’s peak-to-peak ripple current is given OUT 1. External component selection is driven by the load by: requirement and begins with the selection of L and R . SENSE V - V (cid:230) V +V (cid:246) Next, the Power MOSFET and the output diode D1 are IRIPPLE= IN ( )OUT(cid:231) OUT D(cid:247) selected followed by C and C . f L Ł V +V ł IN OUT IN D where f is the operating frequency. Accepting larger values R Selection for Output Current SENSE of I allows the use of low inductances, but results in RIPPLE R is chosen based on the required output current. SENSE higher output voltage ripple and greater core losses. A With the current comparator monitoring the voltage devel- reasonable starting point for setting ripple current is oped across R , the threshold of the comparator SENSE I = 0.4(I ). Remember, the maximum I RIPPLE OUT(MAX) RIPPLE determines the inductor’s peak current. The output cur- occurs at the maximum input voltage. rent the LTC1622 can provide is given by: With Burst Mode operation selected on the LTC1622, the 0.08 I ripple current is normally set such that the inductor IOUT = - RIPPLE current is continuous during the burst periods. Therefore, R 2 SENSE the peak-to-peak ripple current should not exceed: where I is the inductor peak-to-peak ripple current RIPPLE 0.036 (see Inductor Value Calculation section). I £ RIPPLE R A reasonable starting point for setting ripple current is SENSE I = (0.4)(I ). Rearranging the above equation, it RIPPLE OUT This implies a minimum inductance of: becomes: V - V (cid:230) V +V (cid:246) RSENSE = ( )(1 ) for Duty Cycle < 40% LMIN= (cid:230)IN0.03O6UT(cid:246) Ł(cid:231) VOIUNT+VDDł(cid:247) 15 IOUT f(cid:231) (cid:247) Ł R ł SENSE However, for operation that is above 40% duty cycle, slope (Use V = V ) compensation has to be taken into consideration to select IN(MAX) IN the appropriate value to provide the required amount of A smaller value than L could be used in the circuit; MIN current. Using Figure 2, the value of R is: however, the inductor current will not be continuous SENSE during burst periods. SF R = ( )( )( ) SENSE Inductor Core Selection 15 I 100 OUT Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot Inductor Value Calculation afford the core loss found in low cost powdered iron cores, The operating frequency and inductor selection are inter- forcing the use of more expensive ferrite, molypermalloy related in that higher operating frequencies permit the use or Kool Mu® cores. Actual core loss is independent of core of a smaller inductor for the same amount of inductor size for a fixed inductor value, but it is very dependent on ripple current. However, this is at the expense of efficiency inductance selected. As inductance increases, core losses due to an increase in MOSFET gate charge losses. go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will The inductance value also has a direct effect on ripple increase. Ferrite designs have very low core losses and are current. The ripple current, I , decreases with higher RIPPLE inductance or frequency and increases with higher V or IN Kool Mu is a registered trademark of Magnetics, Inc. 7

LTC1622 APPLICATIOUNS INUFORWMATIOUN preferred at high switching frequencies, so design goals In applications where the maximum duty cycle is less than can concentrate on copper loss and preventing saturation. 100% and the LTC1622 is in continuous mode, the R DS(ON) Ferrite core materials saturate “hard,” which means that is governed by: the inductance collapses abruptly when the peak design P current is exceeded. This results in an abrupt increase in RDS(ON)@ ( ) P( ) inductor ripple current and consequently, output voltage DC I 2 1+d p OUT ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low where DC is the maximum operating duty cycle of the loss core material for toroids, but it is more expensive than LTC1622. ferrite. A reasonable compromise from the same manu- When the LTC1622 is operating in continuous mode, the facturer is Kool Mu. Toroids are very space efficient, MOSFET power dissipation is: especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more V +V ( )2( ) difficult. However, new surface mountable designs that do P = OUT D I 1+d p R MOSFET OUT DS(ON) not increase the height significantly are available. VIN+VD ( ) ( )( )( ) 2 +K V I C f Power MOSFET Selection IN OUT RSS An external P-channel power MOSFET must be selected where K is a constant inversely related to gate drive for use with the LTC1622. The main selection criteria for current. Because of the high switching frequency, the the power MOSFET are the threshold voltage V and GS(TH) second term relating to switching loss is important not to the “on” resistance R ,reverse transfer capacitance DS(ON) overlook. The constant K = 3 can be used to estimate the C and total gate charge. RSS contributions of the two terms in the MOSFET dissipation Since the LTC1622 is designed for operation down to low equation. input voltages, a sublogic level threshold MOSFET (R DS(ON) guaranteed at V = 2.5V) is required for applications that Output Diode Selection GS work close to this voltage. When these MOSFETs are used, The catch diode carries load current during the off-time. make sure that the input supply to the LTC1622 is less than The average diode current is therefore dependent on the the absolute maximum MOSFET V rating, typically 8V. GS P-channel switch duty cycle. At high input voltages the The gate drive voltage levels are from ground to V . IN diode conducts most of the time. As V approaches V IN OUT The required minimum RDS(ON) of the MOSFET is gov- the diode conducts only a small fraction of the time. The erned by its allowable power dissipation. For applications most stressful condition for the diode is when the output that may operate the LTC1622 in dropout, i.e., 100% duty is short circuited. Under this condition the diode must cycle, at its worst case the required R is given by: DS(ON) safely handle I at close to 100% duty cycle. Therefore, PEAK it is important to adequately specify the diode peak current P RDS(ON)DC=100% = ( P)2( ) and average power dissipation so as not to exceed the IOUT(MAX) 1+d p diode ratings. Under normal load conditions, the average current con- where P is the allowable power dissipation and d p is the P ducted by the diode is: temperature dependency of R . (1 + d p) is generally DS(ON) given for a MOSFET in the form of a normalized R vs DS(ON) (cid:230) V - V (cid:246) temperature curve, but d p = 0.005/(cid:176) C can be used as an ID=(cid:231) IN OUT(cid:247) IOUT Ł V +V ł approximation for low voltage MOSFETs. IN D 8

LTC1622 APPLICATIOUNS INUFORWMATIOUN The allowable forward voltage drop in the diode is calcu- (cid:230) (cid:246) 1 lated from the maximum short-circuit current as: D VOUT » IRIPPLE(cid:231) ESR+ (cid:247) Ł 8fC ł OUT P V » D where f is the operating frequency, C is the output F OUT ISC(MAX) capacitance and I is the ripple current in the induc- RIPPLE tor. The output ripple is highest at maximum input voltage where P is the allowable power dissipation and will be D since D I increases with input voltage. L determined by efficiency and/or thermal requirements. The choice of using a smaller output capacitance in- A fast switching diode must also be used to optimize creases the output ripple voltage due to the frequency efficiency. Schottky diodes are a good choice for low dependent term, but can be compensated for by using forward drop and fast switching times. Remember to keep capacitors of very low ESR to maintain low ripple voltage. lead length short and observe proper grounding (see The I pin OPTI-LOOP compensation components can be TH Board Layout Checklist) to avoid ringing and increased optimized to provide stable, high performance transient dissipation. response regardless of the output capacitors selected. C and C Selection Manufacturers such as Nichicon, United Chemicon and IN OUT Sanyo should be considered for high performance through- In continuous mode, the source current of the P-channel hole capacitors. The OS-CON semiconductor dielectric MOSFET is a square wave of duty cycle (V + V )/ OUT D capacitor available from Sanyo has the lowest ESR (size) (V + V ). To prevent large voltage transients, a low ESR IN D product of any aluminum electrolytic at a somewhat input capacitor sized for the maximum RMS current must higher price. Once the ESR requirement for C has been be used. The maximum RMS capacitor current is given by: OUT met, the RMS current rating generally far exceeds the [ ( )]1/2 I requirement. V V - V RIPPLE(P-P) OUT IN OUT C Required I » I In surface mount applications, multiple capacitors may IN RMS MAX V IN have to be paralleled to meet the ESR or RMS current This formula has a maximum at V = 2V , where I handling requirements of the application. Aluminum elec- IN OUT RMS = I /2. This simple worst-case condition is commonly trolytic and dry tantalum capacitors are both available in OUT used for design because even significant deviations do not surface mount configurations. In the case of tantalum, it is offer much relief. Note that capacitor manufacturer’s critical that the capacitors are surge tested for use in ripple current ratings are often based on 2000 hours of life. switching power supplies. An excellent choice is the AVX This makes it advisable to further derate the capacitor, or TPS, AVX TPSV and KEMET T510 series of surface mount to choose a capacitor rated at a higher temperature than tantalum, available in case heights ranging from 2mm to required. Several capacitors may be paralleled to meet the 4mm. Other capacitor types include Sanyo OS-CON, Sanyo size or height requirements in the design. Due to the high POSCAP, Nichicon PL series and the Panasonic SP series. operating frequency of the LTC1622, ceramic capacitors can also be used for C . Always consult the manufacturer Low Supply Operation IN if there is any question. Although the LTC1622 can function down to 2V, the The selection of COUT is driven by the required effective maximum allowable output current is reduced when VIN series resistance (ESR). Typically, once the ESR require- decreases below 3V. Figure 3 shows the amount of change ment is satisfied, the capacitance is adequate for filtering. as the supply is reduced down to 2V. Also shown in The output ripple (D VOUT) is approximated by: Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V. Remember the maximum voltage on the I pin defines TH 9

LTC1622 APPLICATIOUNS INUFORWMATIOUN 101 is limiting the efficiency and which change would produce VREF the most improvement. Efficiency can be expressed as: 100 GE (%) 99 VITH Efficiency = 100% – (h 1 + h 2 + h 3 + ...) A LT where h 1, h 2, etc. are the individual losses as a percent- O D V 98 age of input power. E Z LI MA 97 Although all dissipative elements in the circuit produce R O N losses, four main sources usually account for most of the 96 losses in LTC1622 circuits: 1) LTC1622 DC bias current, 95 2) MOSFET gate charge current, 3) I2R losses, 4) voltage 2.0 2.2 2.4 2.6 2.8 3.0 INPUT VOLTAGE (V) drop of the output diode and 5) transition losses. 1622 F03 1. The V current is the DC supply current, given in the IN Figure 3. Line Regulation of V and V REF ITH electrical characteristics, that excludes MOSFET driver and control currents. V current results in a small loss the maximum current sense voltage that sets the maxi- IN which increases with V . mum output current. IN 2. MOSFET gate charge current results from switching Setting Output Voltage the gate capacitance of the power MOSFET. Each time The LTC1622 develops a 0.8V reference voltage between a MOSFET gate is switched from low to high to low the feedback (Pin 3) terminal and ground (see Figure 4). By again, a packet of charge dQ moves from VIN to ground. selecting resistor R1, a constant current is caused to flow The resulting dQ/dt is a current out of VIN which is through R1 and R2 to set the output voltage. The regulated typically much larger than the DC supply current. In output voltage is determined by: continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the (cid:230) (cid:246) R2 VOUT =0.8(cid:231) 1+ (cid:247) MOSFET, inductor and current shunt. In continuous Ł R1ł mode the average output current flows through L but is “chopped” between the P-channel MOSFET in series For most applications, a 30k resistor is suggested for R1. with R and the output diode. The MOSFET R SENSE DS(ON) To prevent stray pickup, an optional 100pF capacitor is plus R multiplied by duty cycle can be summed SENSE suggested across R1 located close to LTC1622. with the resistance of the inductor to obtain I2R losses. 4. The output diode is a major source of power loss at VOUT high currents and gets worse at high input voltages. LTC1622 R2 3 The diode loss is calculated by multiplying the forward VFB 100pF R1 voltage drop times the diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 1622 F04 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load Figure 4. Setting Output Voltage current increases from 0.5A to 2A. Efficiency Considerations 5. Transition losses apply to the external MOSFET and The efficiency of a switching regulator is equal to the increase with higher operating frequencies and input output power divided by the input power times 100%. It is voltages. Transition losses can be estimated from: often useful to analyze individual losses to determine what 10

LTC1622 APPLICATIOUNS INUFORWMATIOUN Transition Loss = 3(V )2I C (f) IN O(MAX) RSS VOUT LTC1622 Other losses including CIN and COUT ESR dissipative R2 + losses, and inductor core losses, generally account for ITH VFB DFB less than 2% total additional loss. R1 Run/Soft-Start Function 1622 F05 The RUN/SS pin is a dual purpose pin that provides the Figure 5. Foldback Current Limiting soft-start function and a means to shut down the LTC1622. Soft-start reduces input surge current from V by gradu- IN Design Example ally increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. Assume the LTC1622 is used in a single lithium-ion An internal 2.5m A current source charges up an external battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to a minimum of capacitor C . When the voltage on the RUN/SS reaches SS 2.7V. Load current requirement is a maximum of 1.5A but 0.7V the LTC1622 begins operating. As the voltage on most of the time it will be on standby mode, requiring only RUN/SS continues to ramp from 0.7V to 1.8V, the internal 2mA. Efficiency at both low and high load current is current limit is also ramped at a proportional linear rate. important. Output voltage is 2.5V. The current limit begins near 0A (at V = 0.7V) and RUN/SS ends at 0.1/RSENSE (VRUN/SS ‡ 1.8V). The output current In the above application, Burst Mode operation is enabled thus ramps up slowly, reducing the starting surge current by connecting Pin 5 to V . IN required from the input power supply. If the RUN/SS has V +V been pulled all the way to ground, there will be a delay Maximum Duty Cycle= OUT D =93% before the current limit starts increasing and is given by: VIN(MIN)+VD t = 2.8 • 105 • C in seconds DELAY SS From Figure 2, SF = 57%. Pulling the RUN/SS pin below 0.4V puts the LTC1622 into Use the curve of Figure 2 since the operating frequency is a low quiescent current shutdown (I < 15m A). Q the free running frequency of the LTC1622. Foldback Current Limiting SF 0.57 R = ( )( )( ) = ( )( ) =0.0253W As described in the Output Diode Selection, the worst- SENSE 15 I 100 15 1.5A OUT case dissipation occurs with a short-circuited output when the diode conducts the current limit value almost In the application, a 0.025W resistor is used. For the continuously. To prevent excessive heating in the diode, inductor, the required value is: foldback current limiting can be added to reduce the current in proportion to the severity of the fault. 4.2- 2.5 (cid:230) 2.5+0.3(cid:246) Foldback current limiting is implemented by adding diode LMIN= (cid:230) 0.036(cid:246) Ł(cid:231) 4.2+0.3ł(cid:247) =1.33m H D (1N4148 or equivalent) between the output and the I 550kHz(cid:231) (cid:247) FB TH Ł 0.025ł pin as shown in Figure 5. In a hard short (V = 0V), the OUT current will be reduced to approximately 50% of the In the application, a 3.9m H inductor is used to reduce maximum output current. inductor ripple current and thus, output voltage ripple. For the selection of the external MOSFET, the R DS(ON) must be guaranteed at 2.5V since the LTC1622 has to work 11

LTC1622 APPLICATIOUNS INUFORWMATIOUN down to 2.7V. Let’s assume that the MOSFET dissipation layout diagram in Figure 6. Check the following in your is to be limited to P = 250mW and its thermal resistance layout: P is 50(cid:176) C/W. Hence the junction temperature at T = 25(cid:176) C A 1. Is the Schottky diode closely connected between ground will be 37.5(cid:176) C and d p = 0.005 (37.5 – 25) = 0.0625. The at (–) lead of C and drain of the external MOSFET? IN required R is then given by: DS(ON) 2. Does the (+) plate of C connect to the sense resistor IN P RDS(ON)@ ( )P( )=0.11W as closely as possible? This capacitor provides AC 2 current to the MOSFET. DC I 1+d p OUT 3. Is the input decoupling capacitor (0.1m F) connected The P-channel MOSFET requirement can be met by an closely between V (Pin 8) and ground (Pin 6)? IN Si6433DQ. 4. Connect the end of R as close to V (Pin 8) as SENSE IN The requirement for the Schottky diode is the most strin- possible. The V pin is the SENSE+ of the current IN gent when V = 0V, i.e., short circuit. With a 0.025W comparator. OUT R resistor, the short-circuit current through the SENSE 5. Is the trace from the SENSE– (Pin 1) to the Sense Schottky is 0.1/0.025 = 4A. An MBRS340T3 Schottky resistor kept short? Does the trace connect close to diode is chosen. With 4A flowing through, the diode is R ? SENSE rated with a forward voltage of 0.4V. Therefore, the worst- case power dissipated by the diode is 1.6W. The addition 6. Keep the switching node, SW, away from sensitive of D (Figure 5) will reduce the diode dissipation to small signal nodes. FB approximately 0.8W. 7. Does the V pin connect directly to the feedback FB The input capacitor requires an RMS current rating of at resistors? The resistive divider R1 and R2 must be least 0.75A at temperature, and COUT will require an ESR connected between the (+) plate of COUT and signal of 0.1W for optimum efficiency. ground. Optional capacitor C1 should be located as close as possible to the LTC1622. PC Board Layout Checklist R1 and R2 should be located as close as possible to the When laying out the printed circuit board, the following LTC1622. R2 should connect to the output as close to checklist should be used to ensure proper operation of the the load as practicable. LTC1622. These items are illustrated graphically in the + VIN RSENSE CIN 1 SENSE– VIN 8 M1 0.1µF 2 7 ITH PDRV L1 LTC1622 SW 3 6 VOUT RITH VFB GND + 4 RUN/ SYNC/ 5 COUT CITH C1 SS MODE CSS QUIET SGND R1 R2 1622 F06 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 6. LTC1622 Layout Diagram (See PC Board Layout Checklist) 12

LTC1622 TYPICAL APPLICATIONUS LTC1622 1.8V/1.5A Regulator with Burst Mode Operation Disabled VIN 47Cµ1F 2.5V TO 8.5V 16V + R2 U1 1 SENSE– VIN 8 0.025Ω 1 8 L1 2 7 2 7 3.3µH VOUT ITH PDRV 1.8V LTC1622 R3 1.5A R101K 3 VFB GND 6 3 6 93.1k +C2 220µF C2230pF 4 RUN/ SYNC/ 5 4 5 R4 6V SS MODE 75k C4 1622 TA01 560pF C1: AVX TPSD476M016R0150 R2: DALE WSL-1206 0.025Ω C2: AVX TPSD227M006R0100 U1: INTERNATIONAL RECTIFIER FETKYTM IRF7422D2 L1: MURATA LQN6C3R3 LTC1622 2.5V/2A Regulator with Burst Mode Operation Enabled VIN 3.3V TO + C1 8.5V 1 SENSE– VIN 8 R0.202Ω 4176µVF · 2 2 7 ITH PDRV M1 L1 3 LTC1622 6 D1 4.7µH VOUT R1 VFB GND 2.5V 10k R3 2A 4 RUN/ + C2 158k SS SYNC/ 5 150µF C3 MODE 6V 220pF · 2 R4 C4 75k 560pF 1622 TA02 C1: AVX TPSD476M016R0150 L1: COILCRAFT D03316-472 C2: SANYO POSCAP 6TPA47M M1: SILICONIX Si3443DV D1: MOTOROLA MBR320T3 R2: DALE WSL-2010 0.02Ω FETKY is a trademark of International Rectifier Corporation. 13

LTC1622 TYPICAL APPLICATIONUS LTC1622 2.5V/3A Regulator with External Frequency Synchronization VIN 3.3V TO R2 + C1 8.5V 1 SENSE– VIN 8 0.01Ω 4176µVF · 2 2 7 ITH PDRV M1 L1 3 LTC1622 6 D1 4.7µH VOUT R1 VFB GND 2.5V 10k R3 3A 4 RUN/ SYNC/ 5 + C2 158k SS MODE 100µF C3 6.3V 220pF C5640pF 16.550VkPH-Pz · 2 R754k C1: AVX TPSD476M016R0150 L1: COILCRAFT D03316-472 1622 TA03 C2: AVX TPSD107M010R0065 M1: SILICONIX Si3443DV D1: MOTOROLA MBR320T3 R2: DALE WSL-2512 0.01Ω Zeta Converter with Foldback Current Limit VIN 2.5V TO D2 R2 + C1 8.5V 1N4818 1 SENSE– VIN 8 0.04Ω 4176µVF · 2 2 7 ITH PDRV Si3441DV L1B LTC1622 6.2µH R471k 34 RVFUBN/ SYGNNCD/ 65 L1A 47µF+ D1 + C1020µF R2332k V3.O3UVT C3 SS MODE 6.2µH 16V 10V 470pF C4 R4 0.1µF 75k 1622 TA04 C1: AVX TPSD476M016R0150 VIN IOUT(MAX) C2: AVX TPSD107M010R0080 3 2 (V) (A) D1: MOTOROLA MBRS320T3 L1B TOP VIEW L1A 2.5 0.45 L1A, L1B: BH ELECTRONICS BH511-1012 3.3 0.70 R2: DALE WSL-1206 0.04Ω 4 •1 5.0 0.95 6.0 1.00 8.4 1.05 14

LTC1622 PACKAGE DESCRIPTIOUN Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 – 0.004* (3.00 – 0.102) 8 7 6 5 0.193 – 0.006 0.118 – 0.004** (4.90 – 0.15) (3.00 – 0.102) 1 2 3 4 0.040 – 0.006 0.034 – 0.004 (1.02 – 0.15) (0.86 – 0.102) 0.007 0° – 6° TYP (0.18) SEATING 0.021 – 0.006 PLANE 0.012 0.006 – 0.004 (0.53 – 0.015) (0.30) (0.15 – 0.102) 0.0256 REF MSOP (MS8) 1098 (0.65) BSC *DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.228 – 0.244 0.150 – 0.157** (5.791 – 6.197) (3.810 – 3.988) 1 2 3 4 0.010 – 0.020 · 45(cid:176) 0.053 – 0.069 (0.254 – 0.508) (1.346 – 1.752) 0.004 – 0.010 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP (0.101 – 0.254) 0.016 – 0.050 0.014 – 0.019 0.050 (0.406 – 1.270) (0.355 – 0.483) (1.270) TYP BSC *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE SO8 1298 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. 15 However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

LTC1622 TYPICAL APPLICATIONU Small Footprint 3.3V/1A Regulator Efficiency vs Load Current VIN 100 R2 + C1 38..35VV TO VIN = 3.5V 1 SENSE– VIN 8 0.025Ω 11C06EµVRFAMIC 90 R101k 234 RSIVTSUFHBN/LTC162SMP2YGDONNRDCDVE/ 765 D1 M12.L21µCH2+ R2332k 31V.AO3UVT EFFICIENCY (%) 8700 VIN = 6V VIN = 4.2V C2230pF C5640pF 476µVF R754k 60 VOUT = 3.3V RSENSE = 0.025Ω 50 1622 TA05 1 10 100 1000 C1: MURATA CERAMIC GRM235Y5V106Z L1: COILCRAFT D01608C-222 C2: SANYO POSCAP 6TPA47M M1: SILICONIX Si3443DY LOAD CURRENT (mA) D1: MOTOROLA MBRS120LT3 R2: DALE WSL-2010 0.025Ω 1622 TA05b Efficiency vs Load Current With LTC1622 Boost Converter 3.3V/2.5A Configured as Boost Converter 100 12 SENSE– VIN 87 + C1010µF C0.61µF R0.2015Ω 3V.I3NV 90 VROSUENT S=E 5=V 0.015Ω ITH PDRV 10V VIN = 3.3V C4730pF 3 VFB LTC1622GND 6 L41.6µH R3 V52V.O5UAT NCY (%) 80 R331k 4 RSSUN/ SMYONDCE/ 5 M1 D1 105k+ 1C2022V0µF EFFICIE 70 C5 C4 R4 · 2 150pF 0.1µF Si6801DQ 20k 60 1622 TA06a 50 C1, C2: SANYO POSCAP TPB SERIES M1: SILICONIX Si3442DV 0.001 0.01 0.1 1 D1: MOTOROLA MBRD835L R2: DALE WS-L2512 0.015Ω L1: SUMIDA CEP123-4R6 LOAD CURRENT (mA) 1622 TA06b RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1147 Series High Efficiency Step-Down Switching Regulator Controllers 100% DC, 3.5V £ V £ 16V, HV Version Has 20V IN IN LT1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators High Frequency, Small Inductor, High Efficiency LTC1436/LTC1436-PLL High Efficiency, Low Noise, Synchronous Step-Down Converters 24-Pin Narrow SSOP, 3.5V £ V £ 36V IN LTC1438/LTC1439 Dual, Low Noise, Synchronous Step-Down Converters Multiple Output Capability, 3.5V £ V £ 36V IN LTC1474/LTC1475 Low Quiescent Current Step-Down DC/DC Converters Monolithic, MSOP, I = 10m A OUT LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 8-Pin N-Channel Drive, 3.5V £ V £ 36V IN LTC1626 Low Voltage, High Efficiency Step-Down DC/DC Converter Monolithic, Constant Off-Time, 2.5V £ V £ 6V IN LTC1627/LTC1707 Low Voltage, Monolithic Synchronous Step-Down Regulator Low Supply Voltage Range: 2.65V to 8V, 0.5A LTC1628 Dual High Efficiency 2-Phase Step-Down Controller Antiphase Drive, 3.5V £ V £ 36V, Protection IN LTC1772 SOT-23 Current Mode Step-Down Controller 6-Lead SOT-23, 2.5V £ V £ 9.8V, 550kHz IN LTC1735 High Efficiency, Low Noise Synchronous Switching Controller Burst Mode Operation, Protection, 3.5V £ V £ 36V IN 16 Linear Technology Corporation sn1622 1622fs LT/TP 1001 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)4 32-1900 l FAX: (408) 434-0507 l w ww.linear-tech.com ª LINEAR TECHNOLOGY CORPORATION 1998