LT3791 60V 4-Switch Synchronous Buck-Boost LED Driver Controller FeaTures DescripTion n 4-Switch Single Inductor Architecture Allows V The LT®3791 is a synchronous 4-switch buck-boost LED IN Above, Below or Equal to V driver controller. The controller can regulate LED current OUT n Synchronous Switching: Up to 98.5% Efficiency up to 52V of LED string with input voltages above, below, n Wide V Range: 4.7V to 60V or equal to the output voltage. The constant-frequency, IN n Wide V Range: 0V to 60V (52V LED) forced-continuous current mode architecture allows its OUT n ±6% LED Current Accuracy: 0V ≤ V < 52V frequency to be adjusted or synchronized from 200kHz to LED n True Color PWM™ and Analog Dimming 700kHz. No top MOSFET refresh switching cycle is needed n LED and Input Current Regulation with Current in buck or boost operation. With 60V input, 60V output Monitor Outputs capability and seamless transitions between operating n No Top MOSFET Refresh in Buck or Boost regions, the LT3791 is ideal for LED driver applications in n V Disconnected From V During Shutdown automotive, industrial, and even battery-powered systems. OUT IN n Open or Shorted LED Fault Protection For new designs we recommend the LT8391: 60V syn- n Capable of 100W or Greater per IC chronous 4-switch, buck-boost LED controller due to its n 38-Lead TSSOP with Exposed Pad numerous performance improvements over the LT3791. applicaTions L, LT, LTC, LTM, Linear Technology and the Linear logo are registered and True Color PWM is a trademark of Analog Devices, Inc. All other trademarks are the property of their respective owners. n Automotive Head Lamps/Running Lamps n General Purpose Lighting Typical applicaTion 98.5% Efficient 100W (33.3V 3A) Buck-Boost LED Driver VIN 2.2µF 15V TO 58V 0.003Ω 100V ×5 VIN INTVCC 1µF 4.7µF 50Ω BST2 Efficiency vs VIN 499k 470nF IIVVIINNNP BTSGT11 0.01.µ1FµF 1M 45×.057VµF 100 EN/UVLO SWI 10µH 98 BOOST OVLO BG1 34.2k BUCK 15.8k INTVCC %) BUCK-BOOST 200k 200k LT3791 SNSP CY ( 96 28k SHORTLED 0.004Ω 0.033Ω LE3DA, P1O0W0WER CIEN OPENLED SNSN EFFI 94 PGND PWM BG2 92 IVINMON SW2 ISMON CLKOUT TGFB2 9010 20 30 40 50 60 VREF ISP INPUT VOLTAGE (V) CTRL 0.1µF ISN 3791 TA01b PWMOUT SS SYNCVC RT SGND 10nF 22nF 86.6k 300kHz 3791 TA01a 3791fc 1 For more information www.linear.com/LT3791

LT3791 absoluTe MaxiMuM raTings pin conFiguraTion (Note 1) Supply Voltages TOP VIEW Input Supply (VIN) .....................................................60V CTRL 1 38 OVLO SW1, SW2 ......................................................–1V to 60V SS 2 37 FB OPENLED, SHORTLED ...............................................15V PWM 3 36 VC EN/UVLO, IVINP, IVINN, ISP, ISN ..............................60V OPENLED 4 35 RT INTV , (BST1-SW1), (BST2-SW2) .............................6V SHORTLED 5 34 SYNC CC TEST2, SYNC, RT, CTRL, OVLO, PWM .......................6V VREF 6 33 CLKOUT IVINMON, ISMON, FB, SS, V , V ............................6V ISMON 7 32 TEST2 C REF IVINMON 8 31 PWMOUT IVINP-IVINN, ISP-ISN, SNSP-SNSN .......................±0.5V EN/UVLO 9 30 SGND SNSP, SNSN ...........................................................±0.3V 39 IVINP 10 29 TEST1 Operating Junction Temperature (Notes 2, 3) SGND IVINN 11 28 SNSN LT3791E/LT3791I ...............................–40°C to 125°C VIN 12 27 SNSP LT3791H ............................................–40°C to 150°C INTVCC 13 26 ISN LT3791MP .........................................–55°C to 150°C TG1 14 25 ISP Storage Temperature Range ..................–65°C to 150°C BST1 15 24 TG2 Lead Temperature (Soldering, 10 sec) ...................300°C SW1 16 23 NC PGND 17 22 BST2 BG1 18 21 SW2 BG2 19 20 PGND FE PACKAGE 38-LEAD PLASTIC TSSOP TJMAX = 150°C, θJA = 28°C/W EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB orDer inForMaTion http://www.linear.com/product/LT3791#orderinfo LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3791EFE#PBF LT3791EFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 125°C LT3791IFE#PBF LT3791IFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 125°C LT3791HFE#PBF LT3791HFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 150°C LT3791MPFE#PBF LT3791MPFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. elecTrical characTerisTics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at T = 25°C (Note 2). V = 12V, V = 12V unless otherwise noted. A IN EN/UVLO PARAMETER CONDITIONS MIN TYP MAX UNITS Input V Operating Voltage 4.7 60 V IN V Shutdown I V = 0V 0.1 1 µA IN Q EN/UVLO V Operating I (Not Switching) FB = 1.3V, R = 59.0k 3.0 4 mA IN Q T 3791fc 2 For more information www.linear.com/LT3791

LT3791 elecTrical characTerisTics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at T = 25°C (Note 2). V = 12V, V = 12V unless otherwise noted. A IN EN/UVLO PARAMETER CONDITIONS MIN TYP MAX UNITS Logic Inputs EN/UVLO Falling Threshold l 1.16 1.2 1.24 V EN/UVLO Rising Hysteresis 15 mV EN/UVLO Input Low Voltage I Drops Below 1µA 0.3 V VIN EN/UVLO Pin Bias Current Low V = 1V 2 3 4 µA EN/UVLO EN/UVLO Pin Bias Current High V = 1.6V 10 100 nA EN/UVLO CTRL Input Bias Current V = 1V 20 50 nA CTRL CTRL Latch-Off Threshold 175 mV OVLO Rising Shutdown Voltage l 2.85 3 3.15 V OVLO Falling Hysteresis 75 mV Regulation V Voltage l 1.96 2.00 2.04 V REF V Line Regulation 4.7V < V < 60V 0.002 0.04 %/V REF IN V Threshold V = 2V 97.5 100 102.5 mV (ISP-ISN) CTRL l 94 100 106 mV V = 1100mV 87 90 93 mV CTRL l 84 90 96 mV V = 700mV 47.5 50 52.5 mV CTRL l 46 50 54 mV V = 300mV 6.5 10 13.5 mV CTRL l 5 10 15 mV ISP Bias Current 110 µA ISN Bias Current 20 µA LED Current Sense Common Mode Range 0 60 V LED Current Sense Amplifier g 890 µS m ISMON Monitor Voltage V = 100mV l 0.96 1 1.04 V (ISP-ISN) Input Current Sense Threshold V 3V ≤ V ≤ 60V l 46.5 50 54 mV (IVINP-IVINN) IVINP IVINP Bias Current 90 µA IVINN Bias Current 20 µA Input Current Sense Common Mode Range 3 60 V Input Current Sense Amplifier g 2.12 mS m IVINMON Monitor Voltage V = 50mV l 0.96 1 1.04 V (IVINP-IVINN) FB Regulation Voltage 1.194 1.2 1.206 V l 1.176 1.2 1.220 V FB Line Regulation 4.7V < V < 60V 0.002 0.025 %/V IN FB Amplifier g 565 µS m FB Pin Input Bias Current FB in Regulation 100 200 nA V Standby Input Bias Current PWM = 0V –20 20 nA C V (V ) Boost l 42 51 60 mV SENSE(MAX) SNSP-SNSN Buck l –56 –47.5 –39 mV Fault SS Pull-Up Current V = 0V 14 µA SS SS Discharge Current 1.4 µA FB Overvoltage Rising Threshold 1.22 1.25 V Open LED Rising Threshold (V ) V = 0V l 1.127 1.15 1.173 V FB (ISP-ISN) 3791fc 3 For more information www.linear.com/LT3791

LT3791 elecTrical characTerisTics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at T = 25°C (Note 2). V = 12V, V = 12V unless otherwise noted. A IN EN/UVLO PARAMETER CONDITIONS MIN TYP MAX UNITS Open LED Falling Threshold (V ) l 1.078 1.1 1.122 V FB Open LED Falling Threshold (V ) V = 1.2V 5 10 15 mV (ISP-ISN) FB Short LED Falling Threshold (V ) 380 400 450 mV FB OPENLED Pin Output Impedance 1.1 2.0 kΩ SHORTLED Pin Output Impedance 1.1 2.0 kΩ SS Latch-Off Threshold 1.75 V SS Reset Threshold 0.2 V Oscillator Switching Frequency R = 147k 190 200 210 kHz T R = 59.0k 380 400 420 kHz T R = 29.1k 665 700 735 kHz T SYNC Frequency 200 700 kHz SYNC Pin Resistance to GND 90 kΩ SYNC Threshold Voltage 0.3 1.5 V Internal V Regulator CC INTV Regulation Voltage 4.8 5 5.2 V CC Dropout (V – INTV ) I = –10mA, V = 5V 240 350 mV IN CC INTVCC IN INTV Undervoltage Lockout 3.1 3.5 3.9 V CC INTV Current Limit V = 4V 67 mA CC INTVCC PWM PWM Threshold Voltage 0.3 1.5 V PWM Pin Resistance to GND 90 kΩ PWMOUT Pull-Up Resistance 10 20 Ω PWMOUT Pull-Down Resistance 5 10 Ω NMOS Drivers TG1, TG2 Gate Driver On-Resistance V – V = 5V BST SW Gate Pull-Up 2.6 Ω Gate Pull-Down 1.7 Ω BG1, BG2 Gate Driver On-Resistance V = 5V INTVCC Gate Pull-Up 3 Ω Gate Pull-Down 1.2 Ω TG Off to BG On Delay C = 3300pF 60 ns L BG Off to TG On Delay C = 3300pF 60 ns L TG1, TG2, t R = 59.0k 240 320 ns OFF(MIN) T Note 1: Stresses beyond those listed under Absolute Maximum Ratings operating junction temperature range. The LT3791MP is guaranteed to may cause permanent damage to the device. Exposure to any Absolute meet performance specifications over the –55°C to 150°C operating Maximum Rating condition for extended periods may affect device junction temperature range. High junction temperatures degrade operating reliability and lifetime. lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C. Note 2: The LT3791E is guaranteed to meet performance from 0°C to 125°C junction temperature. Specification over the -40°C to Note 3: The LT3791 includes overtemperature protection that is intended 125°C operating junction temperature range are assured by design, to protect the device during momentary overload conditions. Junction characterization and correlation with statistical process controls. temperature will exceed the maximum operating junction temperature The LT3791I is guaranteed to meet performance specifications over the when overtemperature protection is active. Continuous operation above –40°C to 125°C operating junction temperature range. The LT3791H is the specified absolute maximum operating junction temperature may guaranteed to meet performance specifications over the –40°C to 150°C impair device reliability. 3791fc 4 For more information www.linear.com/LT3791

LT3791 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A INTV Dropout Voltage INTV Current Limit CC CC vs Current, Temperature INTV Voltage vs Temperature vs Temperature CC 2.5 5.20 90 TA = 150°C TA = 25°C 5.15 80 2.0 TA = –50°C A) 70 5.10 m V-V (V)ININTVCC 11..05 INTV (V)CC545...090055 VVIINN == 6102VV CURRENT LIMIT (C 63450000 C 0.5 4.90 INTV 20 4.85 10 0 4.80 0 0 10 20 30 40 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 LDO CURRENT (mA) TEMPERATURE (°C) TEMPERATURE (°C) 3791 G01 3791 G02 3791 G03 INTV Load Regulation V Voltage vs Temperature V Load Regulation CC REF REF 6.00 2.04 2.20 5.75 2.03 2.15 5.50 2.02 2.10 5.25 2.01 2.05 NTV (V)CC5.00 V (V)REF2.00 V (V)REF2.00 I4.75 1.99 1.95 4.50 1.98 1.90 VIN = 60V 4.25 1.97 VIN = 12V 1.85 VIN = 4.7V 4.00 1.96 1.80 0 10 20 30 40 50 60 70 –50 –25 0 25 50 75 100 125 150 0 50 100 150 200 250 300 350 400 ILOAD (mA) TEMPERATURE (°C) IREF (µA) 3791 G04 3791 G05 3791 G06 V Threshold (ISP-ISN) V Threshold vs V V Threshold vs V vs Temperature (ISP-ISN) CTRL (ISP-ISN) ISP 110 108 108 VIN = 12V 100 106 106 90 80 104 104 mV) 70 mV) 102 mV) 102 V ((ISP-ISN) 546000 V ((ISP-ISN)10908 V ((ISP-ISN)10908 30 96 96 20 VISP = 60V 10 94 94 VISP = 12V VISP = 0V 0 92 92 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 0 10 20 30 40 50 60 –50 –25 0 25 50 75 100 125 150 VCTRL (V) VISP (V) TEMPERATURE (°C) 3791 G07 3791 G08 3791 G09 3791fc 5 For more information www.linear.com/LT3791

LT3791 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A V Threshold vs V ISMON Voltage vs Temperature ISMON Voltage vs V (ISP-ISN) FB (ISP-ISN) 120 1.04 1.0 VIN = 12V 1.03 V(ISP-ISN) = 100mV 0.9 100 0.8 1.02 0.7 mV) 80 V)1.01 V) 0.6 V ((ISP-ISN) 4600 V (ISMON10..0909 V (ISMON 00..45 0.3 0.98 0.2 20 0.97 0.1 0 0.96 0 1.17 1.18 1.19 1.20 1.21 1.22 1.23 –50 –25 0 25 50 75 100 125 150 0 10 20 30 40 50 60 70 80 90 100 VFB (V) TEMPERATURE (°C) V(ISP-ISN) (mV) 3791 G10 3791 G11 3791 G12 V Threshold V Threshold (IVINP-IVINN) (IVINP-IVINN) vs Temperature vs V IVINMON Voltage vs Temperature IVINP 56 52.0 1.04 VIVINP = 12V 54 51.5 1.03 V(IVINP-VINN) = 50mV 51.0 1.02 (mV)N) 5502 VIVINP = 60V (mV)N)50.5 (V)N1.01 V-(IVINPIVIN 4486 VIVINP = 3V V-(IVINPIVIN5409..05 VIVINMO10..0909 49.0 0.98 44 48.5 0.97 42 48.0 0.96 –50 –25 0 25 50 75 100 125 150 0 10 20 30 40 50 60 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) VIVINP (V) TEMPERATURE (°C) 3791 G13 3791 G14 3791 G15 FB Regulation Voltage SHORTLED Threshold V Threshold vs V vs Temperature vs Temperature (IVINP-IVINN) FB 60 1.24 0.500 1.23 0.475 50 1.22 0.450 (mV)P-IVINSN) 3400 V (V)FB11..2201 VOLTAGE (V) 00..440205 FRAILSLIINNGG V(IVIN 20 1.19 FB 0.375 1.18 0.350 10 VIN = 60V 1.17 VIN = 12V 0.325 VIN = 4.7V 0 1.16 0.300 1.17 1.18 1.19 1.20 1.21 1.22 1.23 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 VFB (V) TEMPERATURE (°C) TEMPERATURE (°C) 3791 G16 3791 G17 3791 G18 3791fc 6 For more information www.linear.com/LT3791

LT3791 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A OPENLED Threshold vs Temperature OVLO Threshold vs Temperature Soft-Start Current vs Temperature 1.200 3.3 16 1.175 3.2 14 CHARGING RISING 1.150 V) 3.1 12 GE (V) 1.125 FALLING HOLD ( 3.0 RISING A) 10 VOLTA 1.100 THRES 2.9 FALLING I (µSS 8 B 1.075 O 2.8 6 F L V O 1.050 2.7 4 DISCHARGING 1.025 2.6 2 1.000 2.5 0 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) 3791 G19 3791 G20 3791 G21 Supply Current vs Input Voltage EN/UVLO Pin Current EN/UVLO Threshold Voltage 4.0 8 1.30 VEN/ULO = 1V 1.28 3.5 7 I (mA)Q 21132.....00505 EN/UVLO PIN CURRENT (µA) 46235 EN/UVLO THRESHOLD (V)1111111.......21122122680446 FRAILSLININGG TA = 150°C 0.5 TA = 25°C 1 1.12 TA = –50°C 0 0 1.10 0 10 20 30 40 50 60 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 VIN (V) TEMPERATURE (°C) TEMPERATURE (°C) 3791 G22 3791 G23 3791 G24 Oscillator Frequency TG1, TG2 Minimum On-Time TG1, TG2 Minimum Off-Time vs Temperature vs Temperature vs Temperature 800 100 350 TG2 QUENCY (kHz)657000000 RRTT == 2599..10kk M ON-TIME (ns) 879000 TG1 M OFF-TIME (ns) 232005000 ffSSWW == 240000kkHHzz SWITCHING FRE423100000000 RT = 147k TG1, TG2 MINIMU 64530000 TG1, TG2 MINIMU 11505000 fSW = 700kHz 0 20 0 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (˚C) 3791 G25 3791 G26 3791 G27 3791fc 7 For more information www.linear.com/LT3791

LT3791 Typical perForMance characTerisTics T = 25°C, unless otherwise noted. A V , V UVLO BG1, BG2 Driver On-Resistance TG1, TG2 Driver On-Resistance (BST1-SW1) (BST2-SW2) vs Temperature vs Temperature vs Temperature 3.9 4.5 4.0 3.8 4.0 3.5 V, V (V)(BST1-SW1)(BST2,SW2)33333.....57346 FRAILSLIINNGG BG1, BG2 RESISTANCE (Ω) 323112......005505 PUPLULL-LD-OUWPN TG1, TG2 RESISTANCE (Ω)23112.....00055 PUPLULL-LD-OUWPN 3.2 0.5 0.5 3.1 0 0 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) 3791 G28 3791 G29 3791 G30 PWMOUT On-Resistance V Buck Threshold (SNSP-SNSN) vs Temperature V Voltage vs Duty Cycle vs V C C 14 1.6 60 V(SNSP-SNSN) = 0V 12 1.4 40 Ω) 1.2 BG2 CE ( 10 PULL-UP V) 20 N 1.0 m MOUT RESISTA 864 PULL-DOWN V (V)C 00..86 BG1 V ((SNSP-SNSN)–200 W 0.4 P 2 0.2 –40 0 0 –60 –50 –25 0 25 50 75 100 125 150 0 20 40 60 80 100 0.6 0.8 1.0 1.2 1.4 1.6 1.8 TEMPERATURE (˚C) DUTY CYCLE (%) VC (V) 3791 G31 3791 G32 3791 G33 V Buck Threshold V Boost Threshold V Boost Threshold (SNSP-SNSN) (SNSP-SNSN) (SNSP-SNSN) vs Temperature vs V vs Temperature C 60 60 60 V) 40 VC(MIN) 40 V) 40 VC(MAX) m m HRESHOLD ( 200 (mV)NSN) 200 HRESHOLD ( 200 TP-SNSN)–20 V(SNSP-S––2400 TP-SNSN)––2400 S S N N S S V(–40 VC(MAX) –60 V(–60 VC(MIN) –60 –80 –80 –50 –25 0 25 50 75 100 125 150 0.6 0.8 1.0 1.2 1.4 1.6 1.8 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) VC (V) TEMPERATURE (˚C) 3791 G34 3791 G35 3791 G36 3791fc 8 For more information www.linear.com/LT3791

LT3791 pin FuncTions CTRL (Pin 1): Current Sense Threshold Adjustment Pin for EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate Analog Dimming. Regulating threshold V is 1/10th 1.2V falling threshold with an externally programmable (ISP-ISN) of (V – 200mV). CTRL linear range is from 200mV hysteresis is generated by the external resistor divider CTRL to 1.1V. For V > 1.3V, the current sense threshold is and a 3µA pull-down current. Above the 1.2V (typical) CTRL constant at the full-scale value of 100mV. For 1.1V < V threshold (but below 6V), EN/UVLO input bias current is CTRL < 1.3V, the dependence of the current sense threshold sub-µA. Below the falling threshold, a 3µA pull-down cur- upon V transitions from a linear function to a con- rent is enabled so the user can define the hysteresis with CTRL stant value, reaching 98% of full scale by V = 1.2V. the external resistor selection. An undervoltage condition CTRL Connect CTRL to V for the 100mV default threshold. resets soft-start. Tie to 0.3V, or less, to disable the device REF Force less than 175mV (typical) to stop switching. Do not and reduce V quiescent current below 1µA. IN leave this pin open. IVINP (Pin 10): Positive Input for the Input Current Limit SS (Pin 2): Soft-start reduces the input power sources and Monitor. Input bias current for this pin is typically 90µA. surge current by gradually increasing the controller’s cur- IVINN (Pin 11): Negative Input for the Input Current Limit rent limit. A minimum value of 10nF is recommended on and Monitor. The input bias current for this pin is typically this pin. SS is used as a timer when an open or shorted 20µA. LED condition occurs. A 500k resistor placed from SS to VREF will latch the part off in the event of a fault. If left VIN (Pin 12): Main Input Supply. Bypass this pin to PGND open, a 1.4µA current source pulls down on SS and the with a capacitor. part restarts in a fault. INTV (Pin 13): Internal 5V Regulator Output. The driver CC PWM (Pin 3): A signal low turns off switches, idles switch- and control circuits are powered from this voltage. Bypass ing and disconnects the V pin from all external loads. The this pin to PGND with a minimum 4.7µF ceramic capacitor. C PWMOUT pin follows the PWM pin. PWM has an internal TG1 (Pin 14): Top Gate Drive. Drives the top N-channel 100k pull-down resistor. If not used, connect to INTV . CC MOSFET with a voltage equal to INTV superimposed on CC OPENLED (Pin 4): An open-drain pull-down on OPENLED the switch node voltage SW1. asserts if FB is greater than 1.15V (typical) and V (ISP-ISN) BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin is less than 10mV (typical). To function, the pin requires swings from a diode voltage below INTV up to a diode CC an external pull-up resistor. voltage below V + INTV . IN CC SHORTLED (Pin 5): An open-drain pull-down on SW1 (Pin 16): Switch Node. SW1 pin swings from a diode SHORTLED asserts if FB is less than 400mV (typical). voltage drop below ground up to V . IN To function, the pin requires an external pull-up resistor. PGND (Pins 17, 20): Power Ground. Connect these pins V (Pin 6): Voltage Reference Output Pin, Typically 2V. REF closely to the source of the bottom N-channel MOSFET. This pin drives a resistor divider for the CTRL pin, either BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the for analog dimming or for temperature limit/compensa- bottom N-channel MOSFET between ground and INTV . tion of the LED load. Can supply up to 200µA of current. CC BG2 (Pin 19): Bottom Gate Drive. Drives the gate of the ISMON (Pin 7): Monitor pin that produces a voltage that bottom N-channel MOSFET between ground and INTV . is ten times the voltage V . ISMON will equal 1V CC (ISP-ISN) when V(ISP-ISN) = 100mV. SW2 (Pin 21): Switch Node. SW2 pin swings from a diode voltage drop below ground up to V . IVINMON (Pin 8): Monitor pin that produces a voltage OUT that is twenty times the voltage V(IVINP-IVINN). IVINMON BST2 (Pin 22): Bootstrapped Driver Supply. The BST2 pin will equal 1V when V(IVINP-IVINN) = 50mV. swings from a diode voltage below INTVCC up to a diode voltage below V + INTV . OUT CC 3791fc 9 For more information www.linear.com/LT3791

LT3791 pin FuncTions NC (Pin 23): No Connect Pin. Leave this pin floating. TEST2 (Pin 32): This pin is used for testing purposes only and must be connected to INTV (Pin 13) for the part to CC TG2 (Pin 24): Top Gate Drive. Drives the top N-channel operate properly. MOSFET with a voltage equal to INTV superimposed on CC the switch node voltage SW2. CLKOUT (Pin 33): Clock Output Pin. An in-phase clock is provided at the oscillator frequency to allow for synchro- ISP (Pin 25): Connection Point for the Positive Terminal nizing two devices for extending output power capability. of the Output Current Feedback Resistor. SYNC (Pin 34): External Synchronization Input Pin. This ISN (Pin 26): Connection Point for the Negative Terminal pin is internally terminated to GND with a 90k resistor. of the Output Current Feedback Resistor. The rising edge will be synchronized with the rising edge SNSP (Pin 27): The Positive Input to the Current Sense of the SYNC signal. Comparator. The V pin voltage and controlled offsets C RT (Pin 35): Frequency Set Pin. Place a resistor to GND between the SNSP and SNSN pins, in conjunction with a to set the internal frequency. The range of oscillation is resistor, set the current trip threshold. 200kHz to 700kHz. SNSN (Pin 28): The Negative Input to the Current Sense V (Pin 36): Current Control Threshold and Error Amplifier C Comparator. Compensation Point. The current comparator threshold TEST1 (Pin 29): This pin is used for testing purposes only increases with this control voltage. The voltage ranges and must be connected to SGND for the part to operate from 0.7V to 1.9V. properly. FB (Pin 37): Voltage Loop Feedback Pin. FB is intended SGND (Pin 30, Exposed Pad Pin 39): Signal Ground. for LED protection of an open or shorted LED event. The All small-signal components and compensation should internal transconductance amplifier with output V will C connect to this ground, which should be connected to regulate FB to 1.2V (typical) through the DC/DC converter. PGND at a single point. Solder the exposed pad directly If the FB input is regulating the loop and V < 10mV, (ISP-ISN) to the ground plane. the OPENLED pull-down is asserted. If the FB pin is less than 400mV, the SHORTLED pull-down is asserted. PWMOUT (Pin 31): Buffered Version of PWM Signal for Driving LED Load Disconnect N-Channel MOSFET. The OVLO (Pin 38): Overvoltage Input Pin. This pin is used for PWMOUT pin is driven from INTV . Use of a MOSFET OVLO, if OVLO > 3V then SS is pulled low, the part stops CC with a gate cutoff voltage higher than 1V is recommended. switching and resets. Do not leave this pin open. 3791fc 10 For more information www.linear.com/LT3791

LT3791 block DiagraM 25 26 10 11 12 6 13 ISP ISN IVINP IVINN VIN VREF INTVCC + – + – A2 A1 SHDN_INT REGS A = 10 A = 10 A = 20 A = 24 ISMON 7 ISMON_INT IVINMON_INT TSD + BST1 15 IVINMON A13 TG1 8 A3 14 EN/UVLO – SW1 16 9 – 3µA A4 SHDN_INT SHDN_INT BUCK 1.2V + SS_RESET LOGIC SS LATCH INTVCC PWM A14 BG1 18 PGND 17 A15 BG2 OSC 19 SLOPE_COMP_BOOST RT INTVCC 35 BOOST SYNC LOGIC 34 33 CLKOUT + SW2 21 A16 TG2 A7 24 SLOPE_COMP_BUCK – BST2 22 + SNSP 27 SHORTLED A10 5 + 0.4V – SNSN 28 A5 – IVINMON_INT – A11 – FB 37 FB + 0.2V VREF + + OPENLED A8 14µA 1.2V 4 A6 – + – 1.15V A12 + CTRL 1 R SS RESET + – ISMON_INT Q SS LATCH S A9 – 1.75V INTVCC + 3V PWM A17 A18 3 1.4µA – OVLO 38 SGND PWMOUT 30, 39 SS VC 31 2 36 3791 BD 3791fc 11 For more information www.linear.com/LT3791

LT3791 operaTion The LT3791 is a current mode controller that provides an slowly charged during start-up. This “soft-start” clamping output voltage above, equal to or below the input voltage. prevents abrupt current from being drawn from the input The LTC proprietary topology and control architecture uses power supply. The SS can also be used as a fault timer a current sensing resistor in buck or boost operation. The whenever an open or shorted LED is detected. sensed inductor current is controlled by the voltage on The top MOSFET drivers are biased from floating boot- the V pin, which is the output of the feedback amplifiers C strap capacitors C1 and C2, which are normally recharged A11 and A12. The V pin is controlled by three inputs, one C through an external diode when the top MOSFET is turned input from the output current loop, one input from the off. Schottky diodes across the synchronous switch M4 input current loop, and the third input from the feedback and synchronous switch M2 are not required, but they do loop. Whichever feedback input is higher takes precedence, provide a lower drop during the dead time. The addition forcing the converter into either a constant-current or a of the Schottky diodes typically improves peak efficiency constant-voltage mode. by 1% to 2% at 500kHz. The LT3791 is designed to transition cleanly between the two modes of operation. Current sense amplifier A1 Power Switch Control senses the voltage between the IVINP and IVINN pins and Figure 1 shows a simplified diagram of how the four provides a pre-gain to amplifier A11. When the voltage power switches are connected to the inductor, V , V IN OUT between IVINP and IVINN reaches 50mV, the output of A1 and GND. Figure 2 shows the regions of operation for the provides IVINMON_INT to the inverting input of A11 and LT3791 as a function of duty cycle D. The power switches the converter is in constant-current mode. If the current are properly controlled so the transfer between regions is sense voltage exceeds 50mV, the output of A1 increases continuous. When V approaches V , the buck-boost IN OUT causing the output of A11 to decrease, thus reducing the region is reached. amount of current delivered to the output. In this manner the current sense voltage is regulated to 50mV. VIN VOUT The output current amplifier works similar to the input TG1 M1 M4 TG2 current amplifier but with a 100mV voltage instead of L1 50mV. The output current sense level is also adjustable SW1 SW2 by the CTRL pin. Forcing CTRL to less than 1.2V forces BG1 M2 M3 BG2 ISMON_INT to the same level as CTRL, thus providing current-level control. The output current amplifier provides RSENSE rail-to-rail operation. Similarly if the FB pin goes above 3791 F01 1.2V the output of A11 decreases to reduce the current level and regulate the output (constant-voltage mode). Figure 1. Simplified Diagram of the Output Switches The LT3791 provides monitoring pins IVINMON and ISMON that are proportional to the voltage across the input and DMAX BOOST output current amplifiers respectively. (BG2) M1 ON, M2 OFF BOOST REGION PWM M3, M4 SWITCHES DMIN The main control loop is shut down by pulling the EN/ BOOST BUCK-BOOST REGION 4-SWITCH PWM UVLO pin low. When the EN/UVLO pin is higher than 1.2V, DMAX BUCK an internal 14µA current source charges soft-start capaci- (TG1) BUCK REGION M4 ON, M3 OFF PWM M2, M1 SWITCHES tor CSS at the SS pin. The VC voltage is then clamped a DMIN BUCK 3791 F02 diode voltage higher than the SS voltage while the C is SS Figure 2. Operating Regions vs Duty Cycle 3791fc 12 For more information www.linear.com/LT3791

LT3791 operaTion Buck Region (V > V ) where D is the duty cycle of the buck-boost IN OUT (BUCK-BOOST) switch range: Switch M4 is always on and switch M3 is always off during this mode. At the start of every cycle, synchronous switch D = 8% (BUCK-BOOST) M2 is turned on first. Inductor current is sensed when Figure 3 shows typical buck operation waveforms. If V IN synchronous switch M2 is turned on. After the sensed approaches V , the buck-boost region is reached. OUT inductor current falls below the reference voltage, which is proportional to V , synchronous switch M2 is turned off C Buck-Boost Region (V ~ V ) IN OUT and switch M1 is turned on for the remainder of the cycle. When V is close to V , the controller is in buck-boost Switches M1 and M2 will alternate, behaving like a typical IN OUT operation. Figure 4 and Figure 5 show typical waveforms in synchronous buck regulator. The duty cycle of switch M1 this operation. Every cycle the controller turns on switches increases until the maximum duty cycle of the converter M2 and M4, then M1 and M4 are turned on until 180° later in buck operation reaches D , given by: MAX(BUCK, TG1) when switches M1 and M3 turn on, and then switches D = 100% – D MAX(BUCK,TG1) (BUCK-BOOST) M1 and M4 are turned on for the remainder of the cycle. M2 + M4 M2 + M4 M2 + M4 M1 + M4 M1 + M4 M1 + M4 3791 F03 Figure 3. Buck Operation (V > V ) IN OUT M1 + M4 M1 + M4 M1 + M4 M2 + M4 M2 + M4 M2 + M4 M1+ M3 M1+ M3 M1+ M3 M1 + M4 M1 + M4 M1 + M4 3791 F04 Figure 4. Buck-Boost Operation (V ≤ V ) IN OUT M1 + M4 M1 + M4 M1 + M4 M2 + M4 M2 + M4 M2 + M4 M1 + M3 M1 + M3 M1 + M3 M1 + M4 M1 + M4 M1 + M4 3791 F05 Figure 5. Buck-Boost Operation (V ≥ V ) IN OUT 3791fc 13 For more information www.linear.com/LT3791

LT3791 operaTion Boost Region (V < V ) Low Current Operation IN OUT Switch M1 is always on and synchronous switch M2 is The LT3791 runs in forced continuous mode. In this mode always off in boost operation. Every cycle switch M3 is the controller behaves as a continuous, PWM current turned on first. Inductor current is sensed when synchro- mode synchronous switching regulator. In boost opera- nous switch M3 is turned on. After the sensed inductor tion, switch M1 is always on, switch M3 and synchronous current exceeds the reference voltage which is proportional switch M4 are alternately turned on to maintain the output to V , switch M3 turns off and synchronous switch M4 voltage independent of the direction of inductor current. C is turned on for the remainder of the cycle. Switches M3 In buck operation, synchronous switch M4 is always on, and M4 alternate, behaving like a typical synchronous switch M1 and synchronous switch M2 are alternately boost regulator. turned on to maintain the output voltage independent of the direction of inductor current. In the forced continuous The duty cycle of switch M3 decreases until the minimum mode, the output can source or sink current. duty cycle of the converter in boost operation reaches D , given by: MIN(BOOST,BG2) D = D MIN(BOOST,BG2) (BUCK-BOOST) where D is the duty cycle of the buck-boost (BUCK-BOOST) switch range: D = 8% (BUCK-BOOST) Figure 6 shows typical boost operation waveforms. If V IN approaches V , the buck-boost region is reached. OUT M1 + M3 M1 + M3 M1 + M3 M1 + M4 M1 + M4 M1 + M4 3791 F06 Figure 6. Boost Operation (V < V ) IN OUT 3791fc 14 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion The Typical Application on the front page is a basic LT3791 The rising edge of CLK_OUT corresponds to the rising edge application circuit. External component selection is driven of SYNC thus allowing paralleling converters. The falling by the load requirement, and begins with the selection of edge of CLK_OUT turns on switch M3 and the rising edge R and the inductor value. Next, the power MOSFETs of CLK_OUT turns on switch M2. SENSE are selected. Finally, C and C are selected. This circuit IN OUT can operate up to an input voltage of 60V. Inductor Selection The operating frequency and inductor selection are inter- Programming The Switching Frequency related in that higher operating frequencies allow the use The RT frequency adjust pin allows the user to program the of smaller inductor and capacitor values. The inductor switching frequency from 200kHz to 700kHz to optimize value has a direct effect on ripple current. The maximum efficiency/performance or external component size. Higher inductor current ripple ΔI can be seen in Figure 7. This L frequency operation yields smaller component size but is the maximum ripple that will prevent subharmonic increases switching losses and gate driving current, and oscillation and also regulate with zero load. The ripple may not allow sufficiently high or low duty cycle operation. should be less than this to allow proper operation over Lower frequency operation gives better performance at the all load currents. For a given ripple the inductance terms cost of larger external component size. For an appropriate in continuous mode are as follows: R resistor value see Table 1. An external resistor from T ( ) V • V –V •100 the RT pin to GND is required; do not leave this pin open. OUT IN(MAX) OUT L > BUCK f•I •%Ripple•V LED IN(MAX) Table 1. Switching Frequency vs R Value T ( ) fOSC (kHz) RT (kΩ) VIN(MIN)2• VOUT–VIN(MIN) •100 L > 200 147 BOOST f•I •%Ripple•V 2 LED OUT 300 84.5 400 59.0 where: 500 45.3 f is operating frequency 600 35.7 % ripple is allowable inductor current ripple 700 29.4 V is minimum input voltage IN(MIN) Frequency Synchronization V is maximum input voltage IN(MAX) The LT3791 switching frequency can be synchronized V is output voltage OUT to an external clock using the SYNC pin. Driving SYNC I is current through the LEDs with a 50% duty cycle waveform is always a good choice, LED otherwise maintain the duty cycle between 10% and 90%. 3791fc 15 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion 200 where ΔIL is peak-to-peak inductor ripple current. In buck 180 operation, the maximum average load current is: 160 ⎛47.5mV ΔI ⎞ %) 140 I = + L (X)120 BOOST ∆IL/ OUT(MAX_BUCK) ⎝⎜ R 2 ⎠⎟ MA ISENSE(MAX) LIMIT SENSE SE(100 N SE 80 The maximum current sensing R value for the boost I/I∆L 60 BUCK ∆IL/ operation is: SENSE ISENSE(MAX) LIMIT 40 2•51mV•V 20 IN(MIN) R = SENSE(MAX) 0 2•I •V +ΔI •V 50 55 60 65 70 75 80 85 90 95 100 LED OUT L(BOOST) IN(MIN) BG1, BG2 DUTY CYCLE (%) 3791 F07 The maximum current sensing RSENSE value for the buck operation is: Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle 2•47.5mV R = SENSE(MAX) For high efficiency, choose an inductor with low core 2•I –ΔI LED L(BUCK) loss. Also, the inductor should have low DC resistance to reduce the I2R losses, and must be able to handle the peak The final RSENSE value should be lower than the calculated inductor current without saturating. To minimize radiated RSENSE(MAX) in both the boost and buck operation. A 20% noise, use a shielded inductor. to 30% margin is usually recommended. RSENSE Selection and Maximum Output Current CIN and COUT Selection R is chosen based on the required output current. The In boost operation, input current is continuous. In buck SENSE current comparator threshold sets the peak of the induc- operation, input current is discontinuous. In buck opera- tor current in boost operation and the maximum inductor tion, the selection of input capacitor, CIN, is driven by the valley current in buck operation. In boost operation, the need to filter the input square wave current. Use a low ESR maximum average load current at V is: capacitor sized to handle the maximum RMS current. For IN(MIN) buck operation, the input RMS current is given by: I =⎛ 51mV – ΔIL⎞•VIN(MIN) OUT(MAX_BOOST) ⎜ ⎟ ΔI 2 ⎝RSENSE 2 ⎠ VOUT IRMS= ILED2•D+ L •D 12 3791fc 16 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion The formula has a maximum at V = 2V . Note that Programming V UVLO and OVLO IN OUT IN ripple current ratings from capacitor manufacturers are The falling UVLO value can be accurately set by the resistor often based on only 2000 hours of life which makes it divider R1 and R2. A small 3µA pull-down current is active advisable to derate the capacitor. when the EN/UVLO is below the threshold. The purpose In boost operation, the discontinuous current shifts of this current is to allow the user to program the rising from the input to the output, so C must be capable hysteresis. The following equations should be used to OUT of reducing the output voltage ripple. The effects of ESR determine the resistor values: (equivalent series resistance) and the bulk capacitance R1+R2 must be considered when choosing the right capacitor V – =1.2• IN(UVLO ) R2 for a given output ripple voltage. The steady ripple due to R1+R2 charging and discharging the bulk capacitance is given by: VIN(UVLO+)=3µA•R1+1.215• R2 ( ) I • V –V LED OUT IN(MIN) ΔVRIPPLE(cid:31)(BOOST_CAP)= The rising OVLO value can be accurately set by the resis- C •V •f OUT OUT tor divider R3 and R4. The following equations should be ΔI used to determine the resistor values: ΔV ≈ L RIPPLE(cid:31)(BUCK_CAP) 8•f•C OUT R3+R4 where COUT is the output filter capacitor. VIN(OVLO+)=3• R4 The steady ripple due to the voltage drop across the ESR R3+R4 V – =2.925• IN(OVLO ) is given by: R4 ΔV = I • ESR BOOST(ESR) LED ΔV = I • ESR VIN BUCK(ESR) LED Multiple capacitors placed in parallel may be needed to LT3791 R1 R3 meet the ESR and RMS current handling requirements. OVLO Output capacitors are also used for stability for the LT3791. EN/UVLO A good starting point for output capacitors is seen in the R2 R4 Typical Applications circuits. Ceramic capacitors have 3791 F08 excellent low ESR characteristics but can have a high voltage coefficient and are recommended for applications Figure 8. Resistor Connection to Set V UVLO and IN less than 100W. Capacitors available with low ESR and OVLO Thresholds high ripple current ratings, such as OS-CON and POSCAP may be needed for applications greater than 100W. 3791fc 17 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion Programming LED Current The CTRL pin should not be left open (tie to V if not REF used). The CTRL pin can also be used in conjunction with The LED current is programmed by placing an appropriate a thermistor to provide overtemperature protection for value current sense resistor, R , in series with the LED LED the LED load, or with a resistor divider to V to reduce string. The voltage drop across R is (Kelvin) sensed IN LED output power and switching current when V is low. by the ISP and ISN pins. The CTRL pin should be tied to IN The presence of a time varying differential voltage signal a voltage higher than 1.2V to get the full-scale 100mV (ripple) across ISP and ISN at the switching frequency (typical) threshold across the sense resistor. The CTRL is expected. The amplitude of this signal is increased by pin can also be used to dim the LED current, although high LED load current, low switching frequency and/or a relative accuracy decreases with the decreasing sense smaller value output filter capacitor. Some level of ripple threshold. When the CTRL pin voltage is less than 1V, signal is acceptable: the compensation capacitor on the the LED current is: V pin filters the signal so the average difference between C I = VCTRL–200mV ISP and ISN is regulated to the user-programmed value. LED R •10 Ripple voltage amplitude (peak-to-peak) in excess of LED 20mV should not cause mis-operation, but may lead to When the CTRL pin voltage is between 1.1V and 1.3V noticeable offset between the average value and the user- the LED current varies with VCTRL, but departs from the programmed value. equation above by an increasing amount as V voltage CTRL increases. Ultimately, when V > 1.3V the LED current ISMON CTRL no longer varies. The typical V threshold vs V (ISP-ISN) CTRL The ISMON pin provides a linear indication of the cur- is listed in Table 2. rent flowing through the LEDs. The equation for V ISMON Table 2. V(ISP-ISN) Threshold vs CTRL is V(ISP–ISN) • 10. This pin is suitable for driving an ADC input, however, the output impedance of this pin is 12.5kΩ V (V) V (mV) CTRL (ISP-ISN) so care must be taken not to load this pin. 1.1 90 1.15 94.5 Programming Input Current Limit 1.2 98 1.25 99.5 The LT3791 has a standalone current sense amplifier. It 1.3 100 can be used to limit the input current. The input current limit is calculated by the following equation: When V is higher than 1.3V, the LED current is CTRL 50mV regulated to: I = IN R 100mV IN I = LED R LED 3791fc 18 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion For loop stability a lowpass RC filter is needed. For The output overvoltage threshold can be set by selecting most applications, a 50Ω resistor and 470nF capacitor the values of R5 and R6 (see Figure 9) according to the is sufficient. following equation: Table 3 R5+R6 V =1.25• OUT(OVP) RIN (mΩ) ILIMIT (A) R6 20 2.5 Make sure the expected V during normal operation does 15 3.3 FB not exceed 1.1V: 12 4.2 R6 10 5.0 V • ≤1.1 LED 6 8.3 R5+R6 5 10.0 These equations set the maximum LED string voltage with 4 12.5 open LED protection for the LT3791 to be 52V. 3 16.7 2 25 Dimming Control There are two methods to control the current source for IVINMON dimming using the LT3791. One method uses the CTRL The IVINMON pin provides a linear indication of the current pin to adjust the current regulated in the LEDs. A second flowing through the input. The equation for VIVINMON is method uses the PWM pin to modulate the current source V • 20. This pin is suitable for driving an ADC (IVINP-IVINN) between zero and full current to achieve a precisely pro- input, however, the output impedance of this pin is 12.5kΩ grammed average current. To make PWM dimming more so care must be taken not to load this pin. accurate, the switch demand current is stored on the V C node during the quiescent phase when PWM is low. This Programming Output Overvoltage Threshold for Open feature minimizes recovery time when the PWM signal goes LED Protection high. To further improve the recovery time a disconnect For an LED driver application with small output capacitors, switch may be used in the LED current path to prevent the the output voltage usually overshoots a lot during an open ISP node from discharging during the PWM signal low LED event. Although the 1.2V (typical) FB regulation loop phase. The minimum PWM on- or off-time is affected by tries to regulate the output, the loop is usually too slow to choice of operating frequency and external component prevent the output from overshooting. Once the FB voltage selection. The best overall combination of PWM and hits its overvoltage threshold, 1.25V (typical), the LT3791 analog dimming capabilities is available if the minimum stops switching by turning TG1, TG2 off, and BG1, BG2 PWM pulse is at least six switching cycles and the PWM on. In this way, the minimum overshoot is guaranteed. pulse is synchronized to the SYNC signal. VOUT LT3791 R5 FB R6 3791 F09 Figure 9. Resistor Connection for Open LED Protection 3791fc 19 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion SHORTLED Pin The SS pin is also used as a fault timer. Once an open LED or a shorted LED fault is detected, a 1.4µA pull- The LT3791 provides an open-drain status pin, down current source is activated. With a 500k pull-up SHORTLED, which pulls low when the FB pin is below resistor to V on the SS pin, the LT3791 will latch off 400mV. The only time the FB pin will be below 400mV REF until the EN/UVLO pin is toggled. Without any resistor is during start-up or if the LEDs are shorted. During to V the SS pin enters a hiccup mode operation. The start-up the LT3791 ignores the voltage on the FB pin REF 1.4µA pulls SS down until 0.2V is reached, at which until the soft-start capacitor reaches 1.75V. To prevent point the 14µA pull-up current source turns on. If the false tripping after startup, a large enough soft-start fault condition hasn’t been removed when SS reaches capacitor must be used to allow the output to get up to 1.75V, then the 1.4µA pull-down current source turns on approximately 40% to 50% of the final value. again initiating a new cycle. This will continue until the OPENLED Pin fault is removed. The LT3791 provides an open-drain status pin, OPENLED, Loop Compensation which pulls low when the FB pin is above 1.15V and the voltage across V is less than 10mV. If the open The LT3791 uses an internal transconductance error (ISP-ISN) LED clamp voltage is programmed correctly using the FB amplifier whose VC output compensates the control loop. pin, then the FB pin should never exceed 1.1V when the The external inductor, output capacitor and the comp- LEDs are connected. Therefore, the only way for the FB ensation resistor and capacitor determine the loop pin to exceed 1.15V is for an open LED event to occur. stability. The inductor and output capacitor are chosen based on Soft-Start, Fault Function performance, size and cost. The compensation resis- Soft-start reduces the input power sources’ surge currents tor and capacitor at V are set to optimize control loop C by gradually increasing the controller’s current limit (pro- response and stability. For typical LED applications, a portional to an internally buffered clamped equivalent of 10nF compensation capacitor at V is adequate, and a C VC). The soft-start interval is set by the soft-start capacitor series resistor should always be used to increase the selection according to the following equation slew rate on the V pin to maintain tighter regulation of C 1.2V LED current during fast transients on the input supply of t = •C SS SS the converter. 14µA Make sure C is large enough when there is loading SS during start-up. 3791fc 20 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion Power MOSFET Selections and Efficiency Switch M2 operates in buck operation as the synchronous Considerations rectifier. Its power dissipation at maximum output current is given by: The LT3791 requires four external N-channel power MOS- FETs, two for the top switches (switch M1 and M4, shown in V –V P = IN OUT •I 2•ρ •R Figure 1) and two for the bottom switches (switch M2 and M2(BUCK) V LED T DS(ON) IN M3 shown in Figure 1). Important parameters for the power Switch M3 operates in boost operation as the control MOSFETs are the breakdown voltage, V , threshold BR(DSS) switch. Its power dissipation at maximum current is voltage, V , on-resistance, R , reverse transfer GS(TH) DS(ON) given by: capacitance, C , and maximum current, I . RSS DS(MAX) (V –V )•V The drive voltage is set by the 5V INTVCC supply. Con- P = OUT IN OUT •I 2• •R M3(BOOST) LED T DS(ON) sequently, logic-level threshold MOSFETs must be used V 2 IN in LT3791 applications. If the input voltage is expected I to drop below the 5V, then sub-logic threshold MOSFETs +k•V 3• LED •C •f OUT RSS V should be considered. IN where C is usually specified by the MOSFET manufac- In order to select the power MOSFETs, the power dis- RSS turers. The constant k, which accounts for the loss caused sipated by the device must be known. For switch M1, the by reverse-recovery current, is inversely proportional to maximum power dissipation happens in boost operation, the gate drive current and has an empirical value of 1.7. when it remains on all the time. Its maximum power dis- sipation at maximum output current is given by: For switch M4, the maximum power dissipation happens in boost operation, when its duty cycle is higher than 2 P =⎛ILED•VOUT⎞ •ρ •R 50%. Its maximum power dissipation at maximum output M1(BOOST) ⎜ ⎟ T DS(ON) ⎝ V ⎠ current is given by: IN 2 where ρ is a normalization factor (unity at 25°C) V ⎛I •V ⎞ T P = IN • LED OUT •ρ •R accounting for the significant variation in on-resistance M4(BOOST) VOUT ⎝⎜ VIN ⎠⎟ T DS(ON) with temperature, typically 0.4%/°C as shown in Figure For the same output voltage and current, switch M1 has 10. For a maximum junction temperature of 125°C, using the highest power dissipation and switch M2 has the low- a value of ρ = 1.5 is reasonable. T est power dissipation unless a short occurs at the output. 3791fc 21 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion From a known power dissipated in the power MOSFET, its diode to be effective, the inductance between it and the junction temperature can be obtained using the following synchronous switch must be as small as possible, mandat- formula: ing that these components be placed adjacently. T = T + P • R J A TH(JA) INTV Regulator CC The R to be used in the equation normally includes TH(JA) An internal P-channel low dropout regulator produces 5V the R for the device plus the thermal resistance from TH(JC) at the INTV pin from the V supply pin. INTV powers CC IN CC the case to the ambient temperature (R ). This value TH(JC) the drivers and internal circuitry within the LT3791. The of T can then be compared to the original, assumed value J INTV pin regulator can supply a peak current of 67mA CC used in the iterative calculation process. and must be bypassed to ground with a minimum of 4.7µF ceramic capacitor or low ESR electrolytic capacitor. An additional 0.1µF ceramic capacitor placed directly adjacent 2.0 Ω) to the INTVCC and PGND IC pins is highly recommended. E ( Good bypassing is necessary to supply the high transient C N 1.5 TA current required by MOSFET gate drivers. S SI E R Higher input voltage applications in which large MOSFETs N- 1.0 O D are being driven at high frequencies may cause the maxi- E Z LI mum junction temperature rating for the LT3791 to be A RM 0.5 exceeded. The system supply current is normally dominated O N ρT by the gate charge current. Additional external loading of 0 the INTVCC also needs to be taken into account for the –50 0 50 100 150 power dissipation calculations. Power dissipation for the JUNCTION TEMPERATURE (°C) IC in this case is V • I , and overall efficiency is 3791 F10 IN INTVCC Figure 10. Normalized R vs Temperature lowered. The junction temperature can be estimated by DS(ON) using the equations given Optional Schottky Diode (D3, D4) Selection TJ = TA + (PD • θJA) The Schottky diodes D3 and D4 shown in the Typical Ap- where θJA (in °C/W) is the package thermal impedance. plications section conduct during the dead time between For example, a typical application operating in continuous the conduction of the power MOSFET switches. They current operation might draw 24mA from a 24V supply: are intended to prevent the body diode of synchronous T = 70°C + 24mA • 24V • 28°C/W = 86°C switches M2 and M4 from turning on and storing charge J during the dead time. In particular, D4 significantly reduces To prevent maximum junction temperature from being reverse-recovery current between switch M4 turn-off and exceeded, the input supply current must be checked switch M3 turn-on, which improves converter efficiency operating in continuous mode at maximum V . IN and reduces switch M3 voltage stress. In order for the 3791fc 22 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion Top Gate (TG) MOSFET Driver Supply (C1, D1, C2, D2) 2. Transition loss. This loss arises from the brief amount of time switch M1 or switch M3 spends in the saturated The external bootstrap capacitors C1 and C2 connected region during switch node transitions. It depends upon to the BST1 and BST2 pins supply the gate drive voltage the input voltage, load current, driver strength and for the topside MOSFET switches M1 and M4. When the MOSFET capacitance, among other factors. The loss top MOSFET switch M1 turns on, the switch node SW1 is significant at input voltages above 20V and can be rises to V and the BST1 pin rises to approximately V + IN IN estimated from: INTV . When the bottom MOSFET switch M2 turns on, the CC switch node SW1 drops low and the bootstrap capacitor Transition Loss ≈ 2.7 • V 2 • I • C • f IN OUT RSS C1 is charged through D1 from INTV . When the bottom CC where C is the reverse-transfer capacitance. RSS MOSFET switch M3 turns on, the switch node SW2 drops low and the bootstrap capacitor C2, is charged through D2 3. INTVCC current. This is the sum of the MOSFET driver from INTV . The bootstrap capacitors C1 and C2 need to and control currents. CC store about 100 times the gate charge required by the top 4. C and C loss. The input capacitor has the difficult IN OUT MOSFET switch M1 and M4. In most applications a 0.1µF job of filtering the large RMS input current to the regu- to 0.47µF, X5R or X7R ceramic capacitor is adequate. lator in buck operation. The output capacitor has the difficult job of filtering the large RMS output current Efficiency Considerations in boost operation. Both C and C are required to IN OUT The power efficiency of a switching regulator is equal to have low ESR to minimize the AC I2R loss and sufficient the output power divided by the input power times 100%. capacitance to prevent the RMS current from causing It is often useful to analyze individual losses to determine additional upstream losses in fuses or batteries. what is limiting the efficiency and which change would 5. Other losses. Schottky diode D3 and D4 are respon- produce the most improvement. Although all dissipative sible for conduction losses during dead time and light elements in circuits produce losses, four main sources load conduction periods. Inductor core loss occurs account for most of the losses in LT3791 circuits: predominately at light loads. Switch M3 causes reverse 1. DC I2R losses. These arise from the resistances of the recovery current loss in boost operation. MOSFETs, sensing resistor, inductor and PC board When making adjustments to improve efficiency, the input traces and cause the efficiency to drop at high output current is the best indicator of changes in efficiency. If you currents. make a change and the input current decreases, then the efficiency has increased. If there is no change in the input current, then there is no change in efficiency. 3791fc 23 For more information www.linear.com/LT3791

LT3791 applicaTions inForMaTion PC Board Layout Checklist n The path formed by switch M1, switch M2, D1 and the C capacitor should have short leads and PC trace IN The basic PC board layout requires a dedicated ground lengths. The path formed by switch M3, switch M4, D2 plane layer. Also, for high current, a multilayer board and the C capacitor also should have short leads OUT provides heat sinking for power components. and PC trace lengths. n The PGND ground plane layer should not have any traces n The output capacitor (–) terminals should be connected and it should be as close as possible to the layer with as close as possible to the (–) terminals of the input power MOSFETs. capacitor. n Place C , switch M1, switch M2 and D1 in one compact IN n Connect the top driver bootstrap capacitor, C1, closely area. Place C , switch M3, switch M4 and D2 in one OUT to the BST1 and SW1 pins. Connect the top driver compact area. bootstrap capacitor, C2, closely to the BST2 and SW2 n Use immediate vias to connect the components (includ- pins. ing the LT3791’s SGND and PGND pins) to the ground n Connect the input capacitors, C , and output capacitors, IN plane. Use several large vias for each power component. C , closely to the power MOSFETs. These capaci- OUT n Use planes for V and V to maintain good voltage tors carry the MOSFET AC current in boost and buck IN OUT filtering and to keep power losses low. operation. n Flood all unused areas on all layers with copper. Flooding n Route SNSN and SNSP leads together with minimum with copper will reduce the temperature rise of power PC trace spacing. Avoid sense lines pass through noisy components. Connect the copper areas to any DC net areas, such as switch nodes. Ensure accurate current (V or PGND). sensing with Kelvin connections at the SENSE resistor. IN n Separate the signal and power grounds. All small-signal n Connect the V pin compensation network close to the C components should return to the SGND pin at one point, IC, between V and the signal ground pins. The capaci- C which is then tied to the PGND pin close to the sources tor helps to filter the effects of PCB noise and output of switch M2 and switch M3. voltage ripple voltage from the compensation loop. n Place switch M2 and switch M3 as close to the control- n Connect the INTV bypass capacitor, C , close to the CC VCC ler as possible, keeping the PGND, BG and SW traces IC, between the INTV and the power ground pins. This CC short. capacitor carries the MOSFET drivers’ current peaks. An additional 0.1µF ceramic capacitor placed immediately n Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and next to the INTV and PGND pins can help improve TG2 nodes away from sensitive small-signal nodes. CC noise performance substantially. 3791fc 24 For more information www.linear.com/LT3791

LT3791 Typical applicaTions 98% Efficient 50W (25V 2A) Buck-Boost LED Driver VIN CIN 4.7V TO 58V 2.2µF RIN 100V 0.003Ω C3 VIN INTVCC CVCC ×4 R7 1µF TEST2 D1 D2 4.7µF 50Ω BST2 33R21k 470Cn7F IIVVIINNNP BTSGT11 M1 C0.101.µ1CFµ2F M4 R1M5 C45×.04O7VUµTF SWI EN/UVLO L1 10µH R6 R2 OVLO BG1 M2 M3 44.2k R3 121k INTVCC 1M LT3791 SNSP R9 R10 200k 200k RSENSE RLED R4 SHORTLED 0.004Ω 0.05Ω 54.9k OPENLED SNSN 25V LED SYNC TWO 100Hz PWM PGND 2A SIGNALS 300kHz SYNC BG2 IVINMON SW2 ISMON TG2 CLKOUT FB R11 C8 VREF ISP 1M 0.1µF ISN CTRL PWMOUT R12 TEST1 237k SS VC RT SGND DL11:, CDO2O: NPXERP BHACT94-160W0J-R 10µH M5 CSS CC R8 M1, M2: RENESAS RJK0651DPB 60VDS 10nF 22nF 86.6k M3, M4: RENESAS RJK0451DPB 40VDS 3791 TA02a 300kHz M5: VISHAY Si2318CDS 40VDS Efficiency vs V 100Hz 1000:1 PWM Dimming 100Hz 250:1 PWM Dimming IN (V = 24V) (V = 24V) IN IN 100 98 PWM PWM 96 BOOST BUCK 5V/DIV 5V/DIV 94 BUCK-BOOST Y (%) 92 5A/DILIV1 5A/DILIV1 C N 90 EFFICIE 88 2A/IDLEIVD 2A/IDLEIVD 86 ISMON ISMON 84 1V/DIV 1V/DIV 82 5µs/DIV 3791 TA02c 5µs/DIV 3791 TA02d 80 0 10 20 30 40 50 60 INPUT VOLTAGE (V) 3791 TA02b 3791fc 25 For more information www.linear.com/LT3791

LT3791 package DescripTion Please refer to http://www.linear.com/product/LT3791#packaging for the most recent package drawings. FE Package 38-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1772 Rev C) Exposed Pad Variation AA 4.75 REF 9.60 – 9.80* (.378 – .386) 4.75 REF (.187) 38 20 6.60 ±0.10 2.74 REF 4.50 REF SEE NOTE 4 6.40 2.74 0.315 ±0.05 REF(.252) (.108) BSC 1.05 ±0.10 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT 1 19 1.20 4.30 – 4.50* (.047) (.169 – .177) 0.25 MAX REF 0° – 8° 0.50 0.09 – 0.20 0.50 – 0.75 (.0196) 0.05 – 0.15 (.0035 – .0079) (.020 – .030) BSC (.002 – .006) 0.17 – 0.27 (.0067 – .0106) FE38 (AA) TSSOP REV C 0910 TYP NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 4. RECOMMENDED MINIMUM PCB METAL SIZE 2. DIMENSIONS ARE INMILLIMETERS FOR EXPOSED PAD ATTACHMENT (INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH 3. DRAWING NOT TO SCALE SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3791fc 26 For more information www.linear.com/LT3791

LT3791 revision hisTory REV DATE DESCRIPTION PAGE NUMBER A 08/12 Clarified Features and Description 1 Clarified graph labels/titles 8 Clarified Pin Functions 9,10 Clarified buck-boost function 13 Clarified programming output for overvoltage or open led/overvoltage threshold 19 Clarified Typical Application 25, 28 B 12/13 Clarified TG1, TG2, t parameters 4 OFF(MIN) C 05/17 Clarified last paragraph on the Description 1 Clarified FB pin input bias current limits 3 Clarified TG1, TG2, t limits 4 OFF(MIN) 3791fc 27 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnecFtioorn mof oitrse c iirncfuoitrsm asa dtieosncr wibewdw h.elrineiena wr.icll onmot /inLTfr3in7g9e 1on existing patent rights.

LT3791 Typical applicaTion 98.5% Efficient 100W (33.3V 3A) Buck-Boost LED Driver RIN 0.003Ω VIN CIN 15V TO 58V 2.2µF 100V VIN INTVCC ×5 C3 D1 D2 CVCC R7 1µF 4.7µF 50Ω BST2 C7 IVINN BST1 C0.21µF COUT R1 470nF IVINP TG1 M1 C0.11µF M4 R1M5 45.07VµF 499k SWI ×5 EN/UVLO L1 R6 R2 OVLO BG1 M2 10µH M3 34.2k 15.8k INTVCC R9 R10 LT3791 SNSP R283k 200k 200k SHORTLED R0.S0E0N4SΩE 0.0R3L3EΩD LE3DA, P1O0W0WER OPENLED SNSN PGND PWM BG2 IVINMON SW2 ISMON TG2 CLKOUT FB VREF ISP CTRL C8 ISN 0.1µF PWMOUT SS SYNCVC RT SGND D1, D2: NXP BAT46WJ CSS CC R8 L1: COOPER HC9-100-R 10µH 10nF 22nF 86.6k M1, M2: RENESAS RJK0651DPB 60VDS M5 300kHz M3, M4: RENESAS RJK0451DPB 40VDS M5: VISHAY SI2318CDS 40VDS 3791 TA03 relaTeD parTs PART NUMBER DESCRIPTION COMMENTS LTC®3780 High Efficiency, Synchronous, 4-Switch Buck-Boost V : 4V to 36V, V Range: 0.8V to 30V, I < 55µA, SSOP-24, QFN-32 IN OUT SD Controller Packages LTC3789 High Efficiency, Synchronous, 4-Switch Buck-Boost VIN: 4V to 38V, VOUT Range: 0.8V to 38V, ISD < 40µA, 4mm × 5mm QFN-28, Controller SSOP-28 Packages LT3755/LT3755-1 High Side 60V, 1MHz LED Controller with True Color V : 4.5V to 40V, V Range: 5V to 60V, 3000:1 True Color PWM, Analog, IN OUT LT3755-2 3000:1 PWM Dimming ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3756/LT3756-1 High Side 100V, 1MHz LED Controller with True Color V : 6V to 100V, V Range: 5V to 100V, 3000:1 True Color PWM, Analog, IN OUT LT3756-2 3000:1 PWM Dimming ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3596 60V, 300mA Step-Down LED Driver V : 6V to 60V, V Range: 5V to 55V, 10000:1 True Color PWM, Analog, IN OUT ISD < 1µA, 5mm × 8mm QFN-52 Package LT3743 Synchronous Step-Down 20A LED Driver with V : 5.5V to 36V, V Range: 5.5V to 35V, 3000:1 True Color PWM, Analog, IN OUT Thee-State LED Current Control ISD < 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages 3791fc 28 LT 0517 REV C • PRINTED IN USA www.linear.com/LT3791 For more information www.linear.com/LT3791  LINEAR TECHNOLOGY CORPORATION 2012