LT1373 250kHz Low Supply Current High Efficiency 1.5A Switching Regulator FEATURES DESCRIPTIOU n 1mA I at 250kHz The LT®1373 is a low supply current high frequency Q n Uses Small Inductors: 15m H current mode switching regulator. It can be operated in all n All Surface Mount Components standard switching configurations including boost, buck, n Only 0.6 Square Inch of Board Space flyback, forward, inverting and “Cuk.” A 1.5A high effi- n Low Minimum Supply Voltage: 2.7V ciency switch is included on the die, along with all oscilla- n Constant Frequency Current Mode tor, control and protection circuitry. All functions of the n Current Limited Power Switch: 1.5A LT1373 are integrated into 8-pin SO/PDIP packages. n Regulates Positive or Negative Outputs Compared to the 500kHz LT1372, which draws 4mA of n Shutdown Supply Current: 12m A Typ quiescent current, the LT1373 switches at 250kHz, typi- nEasy External Synchronization cally consumes only 1mA and has higher efficiency. High n 8-Pin SO or PDIP Packages frequency switching allows for small inductors to be used. All surface mount components consume less than 0.6 APPLICATIOU S square inch of board space. New design techniques increase flexibility and maintain n Boost Regulators ease of use. Switching is easily synchronized to an exter- n CCFL Backlight Driver nal logic level source. A logic low on the shutdown pin n Laptop Computer Supplies reduces supply current to 12m A. Unique error amplifier n Multiple Output Flyback Supplies circuitry can regulate positive or negative output voltage n Inverting Supplies while maintaining simple frequency compensation tech- niques. Nonlinear error amplifier transconductance re- duces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external compo- nents during overload conditions. , LTC and LT are registered trademarks of Linear Technology Corporation. TYPICAL APPLICATIOU 12V Output Efficiency 5V-to-12V Boost Converter 100 5V L1* D1 Vf =IN 2 =5 05kVHz 22m H MBRS120T3 VOUT† 90 5 12V R1 VIN 215k %) OFF ON 4 S/S VSW 8 1% + CY ( 80 C4** N + C221m**F LT1373 FB 2 †M22AmXF IOUT EFFICIE 70 GND6, 7 VC1 R242.9k 15Lm1H 0IO.3UAT 60 C0.201m F 1% 22 H 0.35A *SUMIDA CD75-220KC (22m H) OR 50 R3 COILCRAFT D03316-153 (15m H) 1 10 100 1000 5k **AVX TPSD226M025R0200 OUTPUT CURRENT (mA) LT1373 • TA02 1

LT1373 ABSOLUTE W AXIW UW RATIU GS PACKAGE/ORDER IU FORW ATIOU (Note 1) Supply Voltage ....................................................... 30V ORDER PART Switch Voltage NUMBER TOP VIEW LT1373............................................................... 35V VC 1 8 VSW LT1373CN8 LT1373IN8 LT1373HV .......................................................... 42V FB 2 7 GND LT1373HVCN8 LT1373HVIN8 S/S Pin Voltage....................................................... 30V NFB 3 6 GND S LT1373CS8 LT1373IS8 Feedback Pin Voltage (Transient, 10ms) .............. – 10V S/S 4 5 VIN LT1373HVCS8 LT1373HVIS8 Feedback Pin Current........................................... 10mA N8 PACKAGE S8 PACKAGE 8-LEAD PDIP 8-LEAD PLASTIC SO Negative Feedback Pin Voltage S8 PART MARKING (Transient, 10ms)............................................. – 10V TTJJMMAAXX == 112255(cid:176)(cid:176)CC,, qq JJAA == 110200(cid:176)(cid:176)CC//WW ((NS88)) 1373 1373H Operating Junction Temperature Range 1373I 1373HI Commercial........................................ 0(cid:176) C to 125(cid:176) C* Industrial......................................... –40(cid:176) C to 125(cid:176) C Consult factory for Military grade parts. Short Circuit......................................... 0(cid:176) C to 150(cid:176) C *Units shipped prior to Date Code 9552 are rated at 100(cid:176) C maximum operating temperature. Storage Temperature Range................ –65(cid:176) C to 150(cid:176) C Lead Temperature (Soldering, 10 sec)................. 300(cid:176) C ELECTRICAL CHARACTERISTICS The l denotes specifications which apply over the full operating temperature range, otherwise specifications are at T = 25(cid:176) C. A V = 5V, V = 0.6V, V = V , V , S/S and NFB pins open, unless otherwise noted. IN C FB REF SW SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS V Reference Voltage Measured at Feedback Pin 1.230 1.245 1.260 V REF V = 0.8V l 1.225 1.245 1.265 V C I Feedback Input Current V = V 50 150 nA FB FB REF l 275 nA Reference Voltage Line Regulation 2.7V £ V £ 25V, V = 0.8V l 0.01 0.03 %/V IN C V Negative Feedback Reference Voltage Measured at Negative Feedback Pin –2.51 –2.45 –2.39 V NFB Feedback Pin Open, V = 0.8V l –2.55 –2.45 –2.35 V C I Negative Feedback Input Current V = V l –12 –7 –2 m A NFB NFB NFR Negative Feedback Reference Voltage 2.7V £ V £ 25V, V = 0.8V l 0.01 0.05 %/V IN C Line Regulation g Error Amplifier Transconductance D I = – 5m A 250 375 500 m mho m C l 150 600 m mho Error Amplifier Source Current V = V – 150mV, V = 1.5V l 25 50 90 m A FB REF C Error Amplifier Sink Current V = V + 150mV, V = 1.5V l 850 1500 m A FB REF C Error Amplifier Clamp Voltage High Clamp, V = 1V 1.70 1.95 2.30 V FB Low Clamp, V = 1.5V 0.25 0.40 0.52 V FB A Error Amplifier Voltage Gain 250 V/V V V Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V C f Switching Frequency 2.7V £ V £ 25V 225 250 275 kHz IN 0(cid:176) C £ T £ 125(cid:176) C l 210 250 290 kHz J –40(cid:176) C £ T £ 0(cid:176) C (I Grade) 200 290 kHz J Maximum Switch Duty Cycle l 85 95 % Switch Current Limit Blanking Time 340 500 ns 2

LT1373 ELECTRICAL CHARACTERISTICS The l denotes specifications which apply over the full operating temperature range, otherwise specifications are at T = 25(cid:176) C. A V = 5V, V = 0.6V, V = V , V , S/S and NFB pins open, unless otherwise noted. IN C FB REF SW SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS BV Output Switch Breakdown Voltage LT1373 l 35 47 V LT1373HV 0(cid:176) C £ T £ 125(cid:176) C l 42 47 V J –40(cid:176) C £ T £ 0(cid:176) C (I Grade) 40 V J V Output Switch “On” Resistance I = 1A l 0.5 0.85 W SAT SW I Switch Current Limit Duty Cycle = 50% l 1.5 1.9 2.7 A LIM Duty Cycle = 80% (Note 2) l 1.3 1.7 2.5 A D I Supply Current Increase During Switch On-Time 10 20 mA/A IN D I SW Control Voltage to Switch Current 2 A/V Transconductance Minimum Input Voltage l 2.4 2.7 V I Supply Current 2.7V £ V £ 25V l 1 1.5 mA Q IN Shutdown Supply Current 2.7V £ V £ 25V, V £ 0.6V IN S/S 0(cid:176) C £ T £ 125(cid:176) C l 12 30 m A J –40(cid:176) C £ T £ 0(cid:176) C (I Grade) 50 m A J Shutdown Threshold 2.7V £ V £ 25V l 0.6 1.3 2 V IN Shutdown Delay l 5 12 100 m s S/S Pin Input Current 0V £ V £ 5V l –10 15 m A S/S Synchronization Frequency Range l 300 340 kHz Note 1: Absolute Maximum Ratings are those values beyond which the life Note 2: For duty cycles (DC) between 50% and 90%, minimum of the device may be impaired. guaranteed switch current is given by I = 0.667 (2.75 – DC). LIM TYPICAL PERFORW AU CE CHARACTERISTICS Switch Saturation Voltage Switch Current Limit Minimum Input Voltage vs Switch Current vs Duty Cycle vs Temperature 1.0 3.0 3.0 150°C 0.9 CH SATURATION VOLTAGE (V) 000000......874365 100°–C55°C 25°C WITCH CURRENT LIMIT (A) 1212....0055 –55°C2152°5C°C AND INPUT VOLTAGE (V) 2222....4682 WIT 0.2 S 0.5 2.0 S 0.1 0 0 1.8 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 0 10 20 30 40 50 60 70 80 90 100 –50 –25 0 25 50 75 100 125 150 SWITCH CURRENT (A) DUTY CYCLE (%) TEMPERATURE (°C) LT1373 • G01 LT1373 • G02 LT1373 • G03 3

LT1373 TYPICAL PERFORW AU CE CHARACTERISTICS Shutdown Delay and Threshold Minimum Synchronization Error Amplifier Output Current vs Temperature Voltage vs Temperature vs Feedback Pin Voltage 2108 21..08 V)P-P 3.0 fSYNC = 330kHz A)100 mSHUTDOWN DELAY (s) 111126840462 SHUTDDEOLWANYSTHHRUETSDHOOWLND 00011011........26804462 SHUTDOWN THRESHOLD (V) MUM SYNCHRONIZATION VOLTAGE ( 10221.....55050 mROR AMPLIFIER OUTPUT CURRENT (––25725500550 125°C 2–55°5C°C 0 0 MINI 0 ER –75 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 –0.3 –0.2 –0.1 VREF 0.1 TEMPERATURE (°C) TEMPERATURE (°C) FEEDBACK PIN VOLTAGE (V) LT1373 • G04 LT1373 • G05 LT1373 • G06 S/S Pin Input Current Switching Frequency Error Amplifier Transconductance vs Voltage vs Feedback Pin Voltage vs Temperature 5 110 500 4 VIN = 5V AL) 100 gm = DD VI ((VFCB)) C mS/S PIN INPUT CURRENT (A) –––1313022 CHING FREQUENCY (% OF TYPI 79536840000000 mTRANSCONDUCTANCE (mho)321400000000 T –4 WI 20 S –5 10 0 –1 0 1 2 3 4 5 6 7 8 9 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 –50 –25 0 25 50 75 100 125 150 S/S PIN VOLTAGE (V) FEEDBACK PIN VOLTAGE (V) TEMPERATURE (°C) LT1373 • G07 LT1373 • G08 LT1373 • G09 V Pin Threshold and High Feedback Input Current Negative Feedback Input C Clamp Voltage vs Temperature vs Temperature Current vs Temperature 2.4 400 0 2.2 VFB = VREF A) –2 VNFB = VNFR 2.0 VC HIGH CLAMP nA) 350 mENT ( –4 N VOLTAGE (V) 1111....6248 NPUT CURRENT ( 223050000 ACK INPUT CURR––––118026 V PIC 1.0 VC THRESHOLD ACK I 150 EEDB–14 0.8 EDB 100 VE F–16 0.6 FE 50 GATI–18 E N 0.4 0 –20 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) LT1373 • G10 LT1373 • G11 LT1373 • G12 4

LT1373 PIU FUU CTIOU S V (Pin 1): Compensation Pin. The V pin is used for floating. To synchronize switching, drive the S/S pin be- C C frequency compensation, current limiting and soft start. It tween 300kHz and 340kHz. is the output of the error amplifier and the input of the V (Pin 5): Input Supply Pin. Bypass V with 10m F or IN IN current comparator. Loop frequency compensation can be more. The part goes into undervoltage lockout when V IN performed with an RC network connected from the V pin C drops below 2.5V. Undervoltage lockout stops switching to ground. and pulls the V pin low. C FB (Pin 2): The feedback pin is used for positive output GND S (Pin 6): The ground sense pin is a “clean” ground. voltage sensing and oscillator frequency shifting. It is the The internal reference, error amplifier and negative feed- inverting input to the error amplifier. The noninverting back amplifier are referred to the ground sense pin. Con- input of this amplifier is internally tied to a 1.245V nect it to ground. Keep the ground path connection to the reference. Load on the FB pin should not exceed 100m A output resistor divider and the V compensation network C when the NFB pin is used. See Applications Information. free of large ground currents. NFB (Pin 3): The negative feedback pin is used for negative GND (Pin 7): The ground pin is the emitter connection of output voltage sensing. It is connected to the inverting the power switch and has large currents flowing through it. input of the negative feedback amplifier through a 400k It should be connected directly to a good quality ground source resistor. plane. S/S (Pin 4): Shutdown and Synchronization Pin. The S/S V (Pin 8): The switch pin is the collector of the power SW pin is logic level compatible. Shutdown is active low and switch and has large currents flowing through it. Keep the the shutdown threshold is typically 1.3V. For normal traces to the switching components as short as possible to operation, pull the S/S pin high, tie it to V or leave it IN minimize radiation and voltage spikes. BLOCK DIAGRAW VIN SW SHUTDOWN LOW DROPOUT S/S ANTI-SAT DELAY AND RESET 2.3V REG 250kHz SYNC LOGIC DRIVER SWITCH OSC 5:1 FREQUENCY SHIFT + NEGATIVE FEEDBACK 400k NFB –AMP 200k COMP FB – + ERROR CURRENT 0.08W AMP AMP + – 1.245V VC AV » 6 REF GND SENSE GND LT1373 • BD 5

LT1373 OPERATIOU The LT1373 is a current mode switcher. This means that put overshoot on start-up or overload recovery. When switch duty cycle is directly controlled by switch current the feedback voltage exceeds the reference by 40mV, rather than by output voltage. Referring to the Block error amplifier transconductance increases ten times, Diagram, the switch is turned “On” at the start of each which reduces output overshoot. The feedback input also oscillator cycle. It is turned “Off” when switch current invokes oscillator frequency shifting, which helps pro- reaches a predetermined level. Control of output voltage tect components during overload conditions. When the is obtained by using the output of a voltage sensing error feedback voltage drops below 0.6V, the oscillator fre- amplifier to set current trip level. This technique has quency is reduced 5:1. Lower switching frequency allows several advantages. First, it has immediate response to full control of switch current limit by reducing minimum input voltage variations, unlike voltage mode switchers switch duty cycle. which have notoriously poor line transient response. Unique error amplifier circuitry allows the LT1373 to Second, it reduces the 90(cid:176) phase shift at mid-frequencies directly regulate negative output voltages. The negative in the energy storage inductor. This greatly simplifies feedback amplifier’s 400k source resistor is brought out closed-loop frequency compensation under widely vary- for negative output voltage sensing. The NFB pin regulates ing input voltage or output load conditions. Finally, it at –2.45V while the amplifier output internally drives the allows simple pulse-by-pulse current limiting to provide FB pin to 1.245V. This architecture, which uses the same maximum switch protection under output overload or main error amplifier, prevents duplicating functions and short conditions. A low dropout internal regulator pro- maintains ease of use. (Consult Linear Technology Mar- vides a 2.3V supply for all internal circuitry. This low keting for units that can regulate down to –1.25V.) dropout design allows input voltage to vary from 2.7V to 25V with virtually no change in device performance. A The error signal developed at the amplifier output is 250kHz oscillator is the basic clock for all internal timing. brought out externally. This pin (VC) has three different It turns “On” the output switch via the logic and driver functions. It is used for frequency compensation, current circuitry. Special adaptive anti-sat circuitry detects onset limit adjustment and soft starting. During normal regula- of saturation in the power switch and adjusts driver tor operation this pin sits at a voltage between 1V (low current instantaneously to limit switch saturation. This output current) and 1.9V (high output current). The error minimizes driver dissipation and provides very rapid amplifier is a current output (gm) type, so this voltage can turn-off of the switch. be externally clamped for lowering current limit. Like- wise, a capacitor coupled external clamp will provide soft A 1.245V bandgap reference biases the positive input of start. Switch duty cycle goes to zero if the V pin is pulled C the error amplifier. The negative input of the amplifier is below the control pin threshold, placing the LT1373 in an brought out for positive output voltage sensing. The error idle mode. amplifier has nonlinear transconductance to reduce out- APPLICATIOU S IU FORW ATIOU Positive Output Voltage Setting VOUT The LT1373 develops a 1.245V reference (V ) from the ( ) REF R1 FB pin to ground. Output voltage is set by connecting the R1 VOUT = VREF 1 +R2 FB pin to an output resistor divider (Figure 1). The FB pin PFINB R1 = R2(VOUT– 1) bias current represents a small error and can usually be 1.245 R2 ignored for values of R2 up to 25k. The suggested value for VREF LT1373 • F01 R2 is 24.9k. The NFB pin is normally left open for positive output applications. Figure 1. Positive Output Resistor Divider 6

LT1373 APPLICATIOU S IU FORW ATIOU Negative Output Voltage Setting A logic low on the S/S pin activates shutdown, reducing the part’s supply current to 12m A. Typical synchronization The LT1373 develops a –2.45V reference (V ) from the NFR range is from 1.05 and 1.8 times the part’s natural switch- NFB pin to ground. Output voltage is set by connecting the ing frequency, but is only guaranteed between 300kHz and NFB pin to an output resistor divider (Figure 2). The –7m A 340kHz. A 12m s resetable shutdown delay network guar- NFB pin bias current (I ) can cause output voltage errors NFB antees the part will not go into shutdown while receiving and should not be ignored. This has been accounted for in a synchronization signal. the formula in Figure 2. The suggested value for R2 is 2.49k. The FB pin is normally left open for negative output Caution should be used when synchronizing above applications. See Dual Polarity Output Voltage Sensing for 330kHz because at higher sync frequencies the ampli- limitations of FB pin loading when using the NFB pin. tude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of –VOUT subharmonic switching only occurs when the duty cycle ( ) INFB R1 –VOUT = VNFB 1 +RR12 + INFB (R1) of the switch is above 50%. Higher inductor values will NFB tend to eliminate problems. PIN (‰ VO)UT‰ – 2.45 R2 R1 = 2.45 + (7 • 10–6) R2 Thermal Considerations VNFR LT1373 • F02 Care should be taken to ensure that the worst-case input Figure 2. Negative Output Resistor Divider voltage and load current conditions do not cause exces- sive die temperatures. The packages are rated at 120(cid:176) C/W Dual Polarity Output Voltage Sensing for SO (S8) and 130(cid:176) C/W for PDIP (N8). Certain applications benefit from sensing both positive Average supply current (including driver current) is: and negative output voltages. One example is the Dual Output Flyback Converter with Overvoltage Protection I = 1mA + DC (I /60 + I • 0.004) IN SW SW circuit shown in the Typical Applications section. Each I = switch current SW output voltage resistor divider is individually set as de- DC = switch duty cycle scribed above. When both the FB and NFB pins are used, the LT1373 acts to prevent either output from going Switch power dissipation is given by: beyond its set output voltage. For example in this applica- P = (I )2 • R • DC SW SW SW tion, if the positive output were more heavily loaded than R = output switch “On” resistance the negative, the negative output would be greater and SW would regulate at the desired set-point voltage. The posi- Total power dissipation of the die is the sum of supply tive output would sag slightly below its set-point voltage. current times supply voltage plus switch power: This technique prevents either output from going unregu- P = (I • V ) + P lated high at no load. Please note that the load on the FB D(TOTAL) IN IN SW pin should not exceed 100m A when the NFB pin is used. Choosing the Inductor This situation occurs when the resistor dividers are used at both FB and NFB. True load on FB is not the full divider For most applications the inductor will fall in the range of current unless the positive output is shorted to ground. 10m H to 50m H. Lower values are chosen to reduce physical See Dual Output Flyback Converter application. size of the inductor. Higher values allow more output current because they reduce peak current seen by the Shutdown and Synchronization power switch which has a 1.5A limit. Higher values also reduce input ripple voltage, and reduce core loss. The dual function S/S pin provides easy shutdown and synchronization. It is logic level compatible and can be When choosing an inductor you might have to consider pulled high, tied to V or left floating for normal operation. maximum load current, core and copper losses, allowable IN 7

LT1373 APPLICATIOU S IU FORW ATIOU component height, output voltage ripple, EMI, fault cur- inductor gets too hot, wire insulation will melt and cause rent in the inductor, saturation, and of course, cost. The turn-to-turn shorts). Keep in mind that all good things following procedure is suggested as a way of handling like high efficiency, low profile and high temperature these somewhat complicated and conflicting requirements. operation will increase cost, sometimes dramatically. 1. Assume that the average inductor current (for a boost 5. After making an initial choice, consider the secondary converter) is equal to load current times V /V and things like output voltage ripple, second sourcing, etc. OUT IN decide whether or not the inductor must withstand Use the experts in the Linear Technology application continuous overload conditions. If average inductor department if you feel uncertain about the final choice. current at maximum load current is 0.5A, for instance, They have experience with a wide range of inductor a 0.5A inductor may not survive a continuous 1.5A types and can tell you about the latest developments in overload condition. Also, be aware that boost convert- low profile, surface mounting, etc. ers are not short-circuit protected, and that under output short conditions, inductor current is limited only Output Capacitor by the available current of the input supply. The output capacitor is normally chosen by its effective 2. Calculate peak inductor current at full load current to series resistance (ESR), because this is what determines ensure that the inductor will not saturate. Peak current output ripple voltage. At 500kHz, any polarized capacitor can be significantly higher than output current, espe- is essentially resistive. To get low ESR takes volume, so cially with smaller inductors and lighter loads, so don’t physically smaller capacitors have high ESR. The ESR omit this step. Powered iron cores are forgiving be- range for typical LT1373 applications is 0.05W to 0.5W . A cause they saturate softly, whereas ferrite cores satu- typical output capacitor is an AVX type TPS, 22m F at 25V, rate abruptly. Other core materials fall in between with a guaranteed ESR less than 0.2W . This is a “D” size somewhere. The following formula assumes continu- surface mount solid tantalum capacitor. TPS capacitors ous mode operation, but it errors only slightly on the are specially constructed and tested for low ESR, so they high side for discontinuous mode, so it can be used for give the lowest ESR for a given volume. To further reduce all conditions. ESR, multiple output capacitors can be used in parallel. The value in microfarads is not particularly critical and VOUT VIN (VOUT – VIN) values from 22m F to greater than 500m F work well, but you I = I • + PEAK OUT VIN 2(f)(L)(VOUT) cannot cheat mother nature on ESR. If you find a tiny 22m F solid tantalum capacitor, it will have high ESR and output V = minimum input voltage IN ripple voltage will be terrible. Table 1 shows some typical f = 250kHz switching frequency solid tantalum surface mount capacitors. 3. Decide if the design can tolerate an “open” core geom- Table 1. Surface Mount Solid Tantalum Capacitor etry like a rod or barrel, which have high magnetic field ESR and Ripple Current radiation, or whether it needs a closed core like a toroid E CASE SIZE ESR (MAX W ) RIPPLE CURRENT (A) to prevent EMI problems. One would not want an open AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 core next to a magnetic storage media for instance! AVX TAJ 0.7 to 0.9 0.4 This is a tough decision because the rods or barrels are D CASE SIZE temptingly cheap and small, and there are no helpful AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 guidelines to calculate when the magnetic field radia- AVX TAJ 0.9 to 2.0 0.36 to 0.24 tion will be a problem. C CASE SIZE AVX TPS 0.2 (Typ) 0.5 (Typ) 4. Start shopping for an inductor which meets the require- AVX TAJ 1.8 to 3.0 0.22 to 0.17 ments of core shape, peak current (to avoid saturation), B CASE SIZE average current (to limit heating), and fault current, (if the AVX TAJ 2.5 to 10 0.16 to 0.08 8

LT1373 APPLICATIOU S IU FORW ATIOU Many engineers have heard that solid tantalum capacitors aluminum electrolytic capacitors may also be used and are prone to failure if they undergo high surge currents. have a high tolerance to turn-on surges. This is historically true and type TPS capacitors are specially tested for surge capability, but surge ruggedness Ceramic Capacitors is not a critical issue with the output capacitor. Solid Higher value, lower cost ceramic capacitors are now tantalum capacitors fail during very high turn-on surges, becoming available in smaller case sizes. These are tempt- which do not occur at the output of regulators. High ing for switching regulator use because of their very low discharge surges, such as when the regulator output is ESR. Unfortunately, the ESR is so low that it can cause dead shorted, do not harm the capacitors. loop stability problems. Solid tantalum capacitor ESR Single inductor boost regulators have large RMS ripple generates a loop “zero” at 5kHz to 50kHz that is instrumen- current in the output capacitor, which must be rated to tal in giving acceptable loop phase margin. Ceramic ca- handle the current. The formula to calculate this is: pacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. Output Capacitor Ripple Current (RMS) They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. DC Linear Technology plans to issue a Design Note on the use I (RMS) = I 1 – DC RIPPLE OUT of ceramic capacitors in the near future. V –V OUT IN = I OUT V Output Diode IN The suggested output diode (D1) is a 1N5818 Schottky or Input Capacitors its Motorola equivalent, MBR130. It is rated at 1A average The input capacitor of a boost converter is less critical due forward current and 30V reverse voltage. Typical forward to the fact that the input current waveform is triangular, voltage is 0.42V at 1A. The diode conducts current only and does not contain large squarewave currents as is during switch-off time. Peak reverse voltage for boost found in the output capacitor. Capacitors in the range of converters is equal to regulator output voltage. Average 10m F to 100m F with an ESR (effective series resistance) of forward current in normal operation is equal to output 0.3W or less work well up to a full 1.5A switch current. current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for boost con- Frequency Compensation verter is: Loop frequency compensation is performed on the output of the error amplifier (V pin) with a series R network. The C C 0.3(V )(V – V ) I = IN OUT IN main pole is formed by the series capacitor and the output RIPPLE (f)(L)(VOUT) impedance (» 1MW ) of the error amplifier. The pole falls in the range of 5Hz to 30Hz. The series resistor creates a f = 250kHz switching frequency “zero” at 2kHz to 10kHz, which improves loop stability and The input capacitor can see a very high surge current when transient response. A second capacitor, typically one tenth a battery or high capacitance source is connected “live”, the size of the main compensation capacitor, is sometimes and solid tantalum capacitors can fail under this condition. used to reduce the switching frequency ripple on the V C Several manufacturers have developed a line of solid pin. V pin ripple is caused by output voltage ripple C tantalum capacitors specially tested for surge capability attenuated by the output divider and multiplied by the error (AVX TPS series, for instance), but even these units may amplifier. Without the second capacitor, V pin ripple is: C fail if the input voltage approaches the maximum voltage rating of the capacitor. AVX recommends derating capaci- 1.245(VRIPPLE)(gm)(RC) V Pin Ripple = C tor voltage by 2:1 for high surge applications. Ceramic and VOUT 9

LT1373 APPLICATIOU S IU FORW ATIOU V = output ripple (V ) The high speed switching current path is shown schemati- RIPPLE P-P g = error amplifier transconductance (» 375m mho) cally in Figure 3. Minimum lead length in this path is m R = series resistor on V pin essential to ensure clean switching and low EMI. The path C C V = DC output voltage including the switch, output diode and output capacitor is OUT the only one containing nanosecond rise and fall times. To prevent irregular switching, V pin ripple should be C Keep this path as short as possible. kept below 50mV . Worst-case V pin ripple occurs at P-P C maximum output load current and will also be increased if SWITCH L1 NODE poor quality (high ESR) output capacitors are used. The VOUT addition of a 0.001m F capacitor on the V pin reduces C switching frequency ripple to only a few millivolts. A low HIGH value for RC will also reduce VC pin ripple, but loop phase VIN FREQUENCY LOAD CIRCULATING margin may be inadequate. PATH Switch Node Considerations LT1373 • F03 For maximum efficiency, switch rise and fall time are made Figure 3 as short as possible. To prevent radiation and high fre- quency resonance problems, proper layout of the compo- More Help nents connected to the switch node is essential. B field (magnetic) radiation is minimized by keeping output di- For more detailed information on switching regulator ode, switch pin and output bypass capacitor leads as short circuits, please see AN19. Linear Technology also offers a as possible. E field radiation is kept low by minimizing the computer software program, SwitcherCADTM, to assist in length and area of all traces connected to the switch pin. designing switching converters. In addition, our applica- A ground plane should always be used under the switcher tions department is always ready to lend a helping hand. circuitry to prevent interplane coupling. SwitcherCAD is a trademark of Linear Technology Corporation. TYPICAL APPLICATIONUS N Positive-to-Negative Converter with Direct Feedback Dual Output Flyback Converter with Overvoltage Protection VIN R2 R1 2.7V TO 16V 275k 302.6k T1* 1% 1% + OFF ON 4C221Sm/FS LTV1I3N573 VSW 8 DPD1N623K41EM4-B185RAS211•30LDT•1343 + R2.2C45753mkF ––V5VOUT† + C1010m F 2 4.75VV TIN5O 13V P6KE-202A, 3•T1*M5BRS140T+3 C473m FV15OVUT NFB 3 R1%3 OFF ON 4 S/FSB VVISNW 8 1N41486, 7 •48 + C4 VC GND †MAX IOUT 21.%49k LT1373 • 47m F 1 6, 7 IOUT VIN NFB 3 1 –VOUT C2 0.3A 3V R4 –15V 0.01m F 0.5A 5V VC GND MBRS140T3 12.4k R1 0.75A 9V 1% 5k 1 6, 7 *COILTRONICS CTX20-2 (407) 241-7876 C2 R5 LT1373 • TA03 0.01m F 2.49k R3 1% 5k *DALE LPE-4841-100MB (605) 665-9301 LT1373 • TA04 10

LT1373 TYPICAL APPLICATIOU S Low Ripple 5V to –3V “Cuk”† Converter VIN L1* V–3OVUT 5V 2 3 250mA 1• •4 C2 R1 47m F 1k 16V 1% 5 8 + VIN VSW + C1 4 22m F S/S C6 10V 7 LT1373 3 0.1m F GND NFB 6 1 GND S VC C3 D1** 47m F + 16V R4 R2 5k 5.49k C4 1% 0.01m F *SUMIDA CLS62-100L LT1373 • TA05 **MOTOROLA MBR0520LT3 †PATENTS MAY APPLY PACKAGE DESCRIPTIOUN Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.400* (10.160) 0.300 – 0.325 0.045 – 0.065 0.130 – 0.005 MAX (7.620 – 8.255) (1.143 – 1.651) (3.302 – 0.127) 8 7 6 5 0.065 0.255 – 0.015* (1.651) 0.009 – 0.015 TYP (6.477 – 0.381) (0.229 – 0.381) 0.125 (3.175) 0.020 (0.325–+00..003155) 0.100 0.018 – 0M.0I0N3 (0M.5I0N8) 1 2 3 4 8.255+0.889 (2.54) (0.457 – 0.076) N8 1098 –0.381 BSC *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020· 45(cid:176) 0.053 – 0.069 8 7 6 5 (0.254 – 0.508) (1.346 – 1.752) 0.004 – 0.010 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP (0.101 – 0.254) 0.228 – 0.244 0.150 – 0.157** 0.016 – 0.050 0.014 – 0.019 0.050 (5.791 – 6.197) (3.810 – 3.988) (0.406 – 1.270) (0.355 – 0.483) (1.270) TYP BSC *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE SO8 1298 **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD 1 2 3 4 FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. 11 However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen- tation that the interconnection of circuits as described herein will not infringe on existing patent rights.

LT1373 TYPICAL APPLICATIOU S 90% Efficient CCFL Supply Two Li-Ion Cells to 5V SEPIC Conveter 5mA MAX VIN LAMP 4V TO 9V C2 27pF 10 D1 1N4148 T1 TO4 3V.50IVVN + 10m F 5 4 3 0.C11m2F 1 OFF ON 4 S/S VIN5 VSW 8 •L313Am H2.C22m F M•BRSD1130LT3 5VVOUT† 330W Q1 Q2 + C32301mVF GNDLT1373 VCFB 2 L313Bm H R715%2k+ C1030m F 1N5818 6, 7 1 10V R1 R3 2.7V5 .T5OV + L1100m H D1N24148 5kC4 214%.9k 2.2m F 5 0.01m F OFF ON 4 S/S VIN VSW 8 52602kW * 10k C1 = AVX TPSD 336M020R0200 †MIOAUXT IOVUITN LT1373 DIMMING C2 = TOKIN 1E225ZY5U-C203-F 0.45A 4V C3 = AVX TPSD 107M010R0100 0.55A 5V 2 L1 = COILTRONICS CTX33-2, SINGLE 0.65A 7V VFB INDUCTOR WITH TWO WINDINGS 0.72A 9V LT1373 • TA07 GND VC 22k 0.1m F 6, 7 + 1 2m F 1N4148 OPTIONAL REMOTE DIMMING LT1372 • TA06 C1 = WIMA MKP-20 CCFL BACKLIGHT APPLICATION CIRCUITS L1 = COILCRAFT D03316-104 CONTAINED IN THIS DATA SHEET ARE Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001 COVERED BY U.S. PATENT NUMBER 5408162 T1 = COILTRONICS CTX 110609 AND OTHER PATENTS PENDING * = 1% FILM RESISTOR DO NOT SUBSTITUTE COMPONENTS COILTRONICS (407) 241-7876 COILCRAFT (708) 639-6400 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1172 100kHz 1.25A Boost Switching Regulator Also for Flyback, Buck and Inverting Configurations LTC®1265 13V 1.2A Monolithic Buck Converter Includes PMOS Switch On-Chip LT1302 Micropower 2A Boost Converter Converts 2V to 5V at 600mA LT1308A/LT1308B 600kHz 2A Switch DC/DC Converter 5V at 1A from a Single Li-Ion Cell LT1370 500kHz High Efficiency 6A Boost Converter 6A, 0.065W Internal Switch LT1372 500kHz 1.5A Boost Switching Regulator Also Regulates Negative Flyback Outputs LT1374 4.5A, 500kHz Step-Down Converter 4.5A, 0.07W Internal Switch LT1376 500kHz 1.5A Buck Switching Regulator Handles Up to 25V Inputs LT1377 1MHz 1.5A Boost Switching Regulator Only 1MHz Integrated Switching Regulator Available LT1613 1.4MHz Switching Regulator in 5-Lead SOT-23 5V at 200mA from 4.4V Input LT1615 Micropower Step-Up DC/DC in 5-Lead SOT-23 20m A I , 36V, 350mA Switch Q LT1949 600kHz, 1A Switch PWM DC/DC Converter 1.1A, 0.5W , 30V Internal Switch, V as Low as 1.5V IN 12 Linear Technology Corporation sn1373 1373fbs LT/TP 0200 2K REV B • PRINTED IN THE USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)4 32-1900 l FAX: (408) 434-0507 l w ww.linear-tech.com ª LINEAR TECHNOLOGY CORPORATION 1995